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MOTOROLA
SEMICONDUCTOR
APPLICATION NOTE
Order this document
by AN98Q/D
AN980
VHF Narrowband FM Receiver Design
Using the MC3362 and the MC3363
Dual Conversion Receivers
Prepared by: Jon Stiiwell
Ricky Ng
Motorola has developed a series of low power narrow-
band FM dual conversion receivers in monolithic silicon
integrated circuits. The MC3362 and the MC3363 are man¬
ufactured in Motorola's MOSAIC process technology.
This process develops NPN transistors with fy = 4 +
GHz, which allows the MC3362 and the MC3363 to have
excellent very high frequency (VHF) operation with low
power drain. They are ideal for application in cordless
phones, narrowband voice and data receivers, CB and
amateur band radios, radio frequency (RF) security
devices and other applications through 200 MHz.
Features of the MC3362/3 Receiver ICs:
• Broadband RF input frequency capability (to 200 MHz
using internal oscillator, over 450 MHz using external
oscillator)
• Single supply operation from V(X = 2 to 7 Vdc
• Low power consumption dec = 3 mA typical at Vqc
= 2 Vdc)
• Internally biased NPN RF transistor amplifier (MC3363)
• Complete dual conversion circuitry — first mixer and
oscillator included
• First local oscillator (LO) includes buffered output and
varactor diode to allow phase locked-loop (PLL) fre¬
quency synthesis for multichannel operation.
• Buffered second local oscillator output available for
PLL reference input (MC3362)
• Multistage limiter and quadrature detection circuitry
included
• RSSI (Received Signal Strength Indicator) with Carrier
Detect logic included
• Built-in data slicing comparator detects zero crossings
of FSK data transmission
• Inverting operational amplifier included for audio mut¬
ing or active filtering (MC3363)
SCOPE
This application note contains functional descriptions
and applications information pertaining to the various
functional blocks of the MC3362/3 receiver circuits. Four
receiver application circuits are shown. A single channel
receiver and a 10 channel frequency synthesized receiver
designed for the 49 MHz cordless telephone band are
shown. A 256 channel "2 Meter" (144-148 MHz) amateur
band receiver is also shown, including an appropriate PLL
frequency synthesizer design to control the receiver's
local oscillator. Finally, a low cost application featuring
the MC3362 as a single chip manually tunable 162 MHz
weatherband receiver is shown. A directory of external
component manufacturers is included as an appendix.
COMPARISON OF THE MC3362 AND THE MC3363
Figures 1A and IB show the system block diagrams of
MC3362 and MC3363, respectively. The MC3362 and the
MC3363 are made from the same die, but a final metal
mask difference allows different features to be made
available on each. Data pertaining to the common func¬
tional blocks are identical on both circuits.
The MC3363 is a complete VHF dual conversion FM
receiver including RF amplifier, two mixers and oscilla¬
tors, limiting IF amplifier and quadrature detection cir¬
cuitry, received signal strength indicator (RSSI) circuitry,
squelch circuitry and a data shaping comparator for
detecting FM frequency shift keyed (FSK) data transmis¬
sions. Receivers using the MC3363 alone can achieve
better than 0.3 /uV input sensitivity for 12 dB SINAD, from
a 50 O source. The MC3363 comes in a 28-lead plastic
wide SOIC package only.
The MC3362 is optimized for cordless telephone appli¬
cations and as such does not contain the RF preamplifier
or squelch circuitry. In addition, the second local oscil¬
lator contains a buffered output so that it can serve as
the system frequency reference in applications where a
10.240 MHz or 10.245 MHz reference is needed. In gen¬
eral, the MC3362 can be substituted for the MC3363
where:
• A receiver with sensitivity of 0.7 /i.V at the input for
12 dB SINAD is adequate.
• An external RF preamplifier with AGC is desired (such
as MOSFET's 3N211 and MPF211).
• Receiver squelch is not needed.
• Surface mount technology cannot be used. The
MC3362 is available in two 24-lead plastic packages
(DIP and wide SOIC surface mount).
MOSAIC is a trademark of Motorola. Inc.
(M) MOTOROLA
©MOTOROLA INC.. 1988
MOTOROLA
2
AN980
1ST MIXER
2NO MIXER
Figure IB. MC3363 internal Block Diagram
MOTOROLA
FEDERAL REGULATIONS,
RECOMMENDED STANDARDS
Radios built for certain VHF and UHF bands may qualify
under the FCC Code of Federal Regulations Title 47, Part
15, for use by unlicensed operators. It is important to
know the federal regulations concerning a particular fre¬
quency channel or band of channels before a receiver or
transmitter circuit is designed. Contact the FCC/Govern-
ment Printing Office to order a copy of the Code of Fed¬
eral Regulations, Title 47, Parts 0-20 which contains Part
15, before designing a radio receiver or transmitter for
unlicensed utility applications.
Professional (landmobile) radios come under another
part (Part 90) of the Title 47 code. There are a set of
standards, published by the Electronic Industries Asso¬
ciation, which dictate recommended operating specifi¬
cations for two way communication equipment. These
standards provide useful information about radio per¬
formance, terminology and measurement techniques
and are useful even if professional radios are not a
designer's primary goal. Contact the EIA at
(202) 457-4300 to order the standards listed below. The
FCC/GPO can be reached at (202)275-2054 or
(213) 894-5841. The pertinent documents are:
Number
FCC Title 47,
Part 15
FCC Title 47,
Part 90
RS-204-C
EIA-152-B
EIA-316-B
Description
Code of Federal
Regulations
Code of Federal
Regulations
EIA Recommended
Standard
EIA Recommended
Standard
EIA Recommended
Standard
Parts Referenced
Radio Frequency
Devices
Landmobile Radios
FM/PM Receiver
Standards
FM/PM Transmitter
Standards
Test Conditions,
Radio Standards
REFERENCE LITERATURE
The following Motorola literature may be useful when
designing with the MC3362/3 receivers:
Number
Description
Parts Referenced
DL128, Rev. 2
Linear and
MC3362, MC3363,
Interface Device
MC34119, MC2831A,
Data
MC2833, MCI3060,
MC33171
DL130
CMOS/NMOS
MCI451XX CMOS
Special
Functions Data
PLL's
DL122
MECL Device
MC12XXX ECL
Data
Prescalers
DL126
Small Signal
Transistor Data
3N211, MPF211
COMPANION DEVICES
e The MC2831A and the MC2833 low power FM trans¬
mitter ICs provide all essential functions for cordless
telephone and general transmitter and oscillator appli¬
cations through 60 MHz (MC2831A) and 200 MHz
(MC2833, using internal very high frequency [VHFJ
transistors as frequency multipliers).
• The MC34119 low power audio amplifier with differ¬
ential outputs provides efficient power transfer and
eliminates the need for the typical large audio coupling
capacitor.
• The MCI3060 Mini-Watt audio amplifier (for higher
powered audio output).
• The MC33171 low power single supply operational
amplifier for use as an RSSI buffer or active integrator.
• The MC14516X series of dual PLL frequency synthes¬
izers for development of 10 channel cordless telephone
band transceivers.
• The MC12XXX series of ECL prescalers and
• MC1451XX series of CMOS Frequency Synthesizers for
development of VHF "high band" radios to 200 MHz
• The MCI45442/3 single chip 300 baud modems which
allow audio frequency shift keyed (AFSK) RF modem
design for very reliable data transmission.
• The 3N211 and MPF211 dual gate MOSFET's for
MC3362 RF preamplification with AGC capability.
BLOCK DESCRIPTION
RF Amplifier (MC3363 only)
The MC3363 contains an internal NPN bipolar RF ampli¬
fier transistor. The base of the transistor is biased inter¬
nally to approximately 0.8 Vdc, which simplifies
common-emitter amplifier design. Grounding the emitter
yields an emitter current lg = 1.5 mA and voltage gain
Ay = 20 dB with a collector load R(_ = 1 kft.
Emitter degeneration resistors can be added to lower
current drain, with Rg decoupling used to preserve the
gain. With the emitter grounded the input at Pin 2 looks
like 180 ft in parallel with 20 pF at 50 MHz. The noise
figure at 50 MHz and unity gain frequency (fy) of the NPN
transistor are approximately 2 dB and 3 GHz, respectively,
at lg = 1.5 mA. The collector load can be resistive, as
shown in Figure 10, ortuned as shown in Figure 14. When
both input and output are tuned and/or impedance
matched care must be taken to prevent unwanted oscil¬
lations — this is why the 2 kft resistor is included in the
collector load of Figure 14.
First Mixer
The first mixer is a doubly balanced multiplier, driven
directly from the RF input and from the first local oscil¬
lator via a cascode amplifier. It is used to convert the RF
input frequency down to the first IF of 10.7 MHz. The
input admittance seen at either RF input pin is 670 ohms
in parallel with 7 pF at 50 MHz; that is, Rp = 670 fi and
Cp = 7 pF. The series equivalent impedance at 50 MHz
is Rs = 210 ft and Cs = 10.2 pF. The first mixer's input
is differential, but can be driven single-ended with no
loss in system gain. If a single-ended input is used, be
sure to AC ground the unused pin. This can be done with
a bypass capacitor to the negative rail (Vgg) or by con¬
necting the pin directly to the Vqq supply.
The isolation of the mixer is shown in Table 1, and of
particular value in many applications will be the strong
attenuation (41 dB) of the local oscillator at the mixer
input. The isolation is due to the fully balanced mixer
configuration used and helps to reduce LO radiation at
the receiver's antenna.
Table 1. First Mixer Isolation Level (in dB) at:
Signal LO Tank Mixer Out (IF) Mixer In (RF)
LO 0 -17 -41
RF -16 -9 0
IF -29 0 <-40
MOTOROLA
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AN980
Initial Alignment of LC Tank
NOTES:
1. The varactor control pin controls the net capacitance across the
local oscillator tank pins. The net capacitance will be approximately
20-25 pF when this pin is left open and 10-15 pF when connected
to Vqc> depending upon strays and the V^c value used. If the V^c
supply is regulated, connect the varactor control pin to VCC :
otherwise, bypass via a 0.01 ju.F capacitor to ground.
2. Connect the oscillator as shown in the left hand drawing (Initial
Alignment) and adjust LI so that the tank resonates approximately
2-3 MHz below the crystal frequency. The frequency should be
checked by examining the buffered output using a high impedance
probe or by some sort of inductive pickup which will not push the
oscillator off frequency. The frequency should be:
^ V(Ll +• L2) Cyaractor
3. Break the L1-L2 connection and add XI and R2. Verify that the LO
operates at the desired frequency by applying an RF input of
Final Configuration
10.7 MHz above or below the crystal frequency and checking for
receiver quieting. Make sure that there is proper LO amplitude. Also
there should be approximately 200 mVpp seen at either LO tank pin.
The buffered LO output should yield 200-600 mVpp depending on
the Vcc value used.
4. The R2 resistor must be included or else the oscillator will latch.
5. This method has been proven effective up to 65 MHz using 3rd
overtone crystals, but has proven unreliable at higher frequencies
(usually using 5th and 7th overtone types). For higher frequency
operation on a single channel, a signal can be injected into the local
oscillator port. (See the "First Mixer and Oscillator" section.)
6. Component values: The R1 resistors should be 10 to 50 kO and are
included in order to add some current and gain to the local
oscillator. LI should equal L2 in nominal value and a fixed value
might be used but startup with different crystals might be degraded.
R2 should be 300 to 1500 O. XI should be 3rd overtone, series
mode resonant (no load capacitance specified).
Figure 2. Running the MC3362/3 First Local Oscillator
on a Single Channel Under Crystal Control
The open circuit conversion voltage gain of the first
mixer is typically 24 dB, flat to 7 MHz. internal rolloff is
provided above 7 MHz to suppress RF and LO signals
and spurious products sent on to the second mixer. The
gain at 10.7 MHz is typically 18 dB. The output circuit is
an emitter follower which is impedance-matched to 330
ohms to drive 10.7 MHz ceramic filters which typically
have 330 ohm input and output impedances. For appli¬
cations which require a high impedance crystal filters,
impedance matching will likely need to be added at the
first mixer's output to preserve the filter's response.
First Local Oscillator and Varactor Diodes
Associated with the first mixer is the first local oscillator
(LO). ft is a complete voltage controlled oscillator and
only requires an external LC tank circuit (no external var¬
actor diode). For multichannel applications, the oscillator
includes varactor tuning and a buffered output suitable
for interfacing to a PLL frequency synthesizer. This is the
approach used in the receivers of Figures 10 and 11. The
maximum oscillation frequency obtained has been
approximately 190 MHz, achieved by injecting extra cur¬
rent into the oscillator. To inject current into the local
oscillator, connect pull-up resistors of 10-50 kfl from Vcc
to each LO tank pin. The LO buffered output varies from
400 mVpp to 1100 mVpp with supply voltage and the
output waveform appears best with Rpd = 3 kfl, as
shown in Figure 3.
There are internal varactor diodes which have capac¬
itance which appears across the local oscillator tank pins.
The internal capacitance can range from 10 to 25 pF
depending on the control voltage applied to the varactor
control pin (MC3362 Pin 23, MC3363 Pin 27). The capac¬
itance is maximum when the voltage applied is at the
minimum (0.7 V) value. Applying voltages greater than
Vcc and lower than 0.7 V to the varactor control pin can
cause the oscillator to stop.
The first local oscillator can be crystal controlled to run
on a single channel. The procedure of Figure 2 shows
how to do this for applications through 65 MHz. The
receiver of Figure 10 uses this approach.
■£ 1 • Vcc “ 5V f§§
■■■ • f 0 * 39 MHz ■pi
A A
\ 1 1
f
\ f
\ ! \ I
\ /
\ /
\ !
\ 1
\ 1 \ 1
\ 1 \!
\l
\!
\l \l
V
V
V V
20 CbiC
10ns
Figure 3. First Local Oscillator Buffered Output
AN980
MOTOROLA
5
A third application of the local oscillator is to drive it
from an external source. This is recommended for appli¬
cations from 75 MHz to 200 MHz and beyond which do
not require PLL frequency synthesis. The inputs are dif¬
ferential and they must be driven using a wideband RF
transformer or balun. The input voltage seen at either
tank pin should be roughly 100 mVrms to ensure proper
operation of the mixer and care should be taken so that
any inductance present at the LO tank pins does not res¬
onate with the internal varactor capacitance (a small val¬
ued resistor of 50-100 ft should ensure this does not
occur). Using this approach, no loss in mixer gain is seen
until the RF and LO inputs are taken over 450 MHz. The
RF and LO inputs should be run with a 10.7 MHz differ¬
ence in frequency to accommodate the first IF bandwidth,
so image frequency considerations (preselector filter
quality) may limit the maximum RF input frequency to
less than 450 MHz.
Second Mixer and Second Local Oscillator
After the 10.7 MHz IF signal is filtered using a ceramic
filter, it is applied to the second mixer input. The second
mixer is also doubly balanced to reduce spurious
responses and typically is used to convert the 10.7 MHz
IF down to 455 kHz for application to the limiting amplifier
and detection circuitry. In the typical low cost application,
the mixer is driven single-endedly from a ceramic filter,
with one of the mixer inputs bypassed directly to the
supply. The open circuit conversion voltage gain is typ¬
ically 25 dB. For applications which require a high imped¬
ance crystal filter, impedance matching will likely need
to be added at the second mixer input to preserve the
filter response. The second mixer output is rolled off
above 500 kHz, to reduce spurious response and idle
noise.
The second local oscillator is a Colpitts type which is
typically run under crystal control. The crystal used is
specified for fundamental mode operation, calibrated for
parallel resonance with a load capacitance of 30-40 pF.
The typical waveform seen at the base is shown in Figure
4. The oscillator can be run at 10.240 MHz or 10.245 MHz,
depending on the first local oscillator frequency desired.
The MC3362 second local oscillator has a buffered output
available which can be used to drive the reference fre¬
quency input of a PLL synthesizer or a prescaler. An exter¬
nal local oscillator signal can be injected into the local
oscillator's base, with the emitter pin left open. The signal
should be sinusoidal and should be approximately 300
mVpp to 500 mVpp in level.
The output admittance of the second mixer at 500 kHz
is 1500 ft in parallel with 50 pF; that is, Rp = 1500 ft and
Cp = 50 pF. The series equivalent impedance is Rs =
1420 ft and Cs = 1065 pF. This impedance matches the
typical input impedance of standard 455 kHz ceramic fil¬
ters, which have 1500-2000 ft typical input and output
impedances.
Limiting IF Amplifier and Quadrature Detector
The 455 kHz IF signal is applied to the limiting IF ampli¬
fier, where it is amplified and limited before application
to the quadrature detection circuitry. The limiting IF
amplifier input has an input impedance of approximately
1.5 kft, which provides good power transfer from 1.5 kft
ceramic filters. The limiting IF circuitry has 10 juV input
sensitivity for —3 dB limiting, flat to 1 MHz. In order to
preserve overall power supply current drain, the limiting
IF and the receiver in general are not designed for wide¬
band applications.
The coupling capacitor from limiter output to quadra¬
ture tank and detector input is provided internally and its
value is 5 pF. The 455 kHz oscillator circuit is typically
built around an LC tank circuit, with Cp = 180 pF, Lp ■=
680 iiH. Typical ceramic resonators can not be driven
from the quadrature tank pin. A waveform like that of
Figure 5 should appear at the quadrature tank pin during
periods of full receiver quieting and no modulation.
Meter Drive (RSSI)
The amplitude of the RF input signal at the appropriate
frequency is monitored by meter drive circuitry. This cir¬
cuitry detects the amount of limiting in the limiting IF
amplifier and produces a linear change in current (nom¬
inally 0.1 fxA) at the meter drive pin for each decibel of
change in the RF input. The meter drive circuitry is fairly
r\
i \
/ \
• Vcc = 5V
• f 0 =■ 10.245 MH:
• Waveform is
observed at
oscillator base
r\
[ \
i \
\ /
V
\J
• V CC = 5V
• f 0 - 455 kHz
into limiting IF
25aC £ 500ns
Figure 4. Second Local Oscillator Waveform Figure 5. Quadrature Tank Pin Waveform Under
Strong Received Signal Condition
AN98Q
MOTOROLA
6
linear for input signal levels over a 60 dB range. This
output can be used as a meter drive or Received Signal
Strength Indicator (RSSI) and needs to be bufffered. In
order to provide a linear, wide ranging RSSI output volt¬
age, three things must be accomplished:
1. The Meter Drive pin (MC3362 Pin 10, MC3363 Pin 12)
should be clamped to within VgE/2 (approximately 300
mV) of the MC3362/3 supply voltage, or loading of the
Meter Drive's current source will occur. The carrier
detect output Is disabled (high output) when the
Meter Drive pin is clamped in this manner. There are
diodes present at the Meter Drive pin which can inter¬
fere with the Meter Drive. (See Figure 6 for a schematic
representation.) With these diodes present the voltage
swing possible at the Meter Drive pin is limited to a
diode drop above and below the Vcc supply.
2. Some type of current to voltage conversion must take
place. The RSSI output is typically 4 to 12 /jlA .
3. Negative feedback must be provided in the output
buffer to counteract buffer amplifier gain variations.
Some method of output level adjustment may be
desirable.
Vcc
Figure 6. Schematic Representation of Meter Drive
"Parasitic Circuits"
Carrier Detect
Another configuration for the meter drive and carrier
detect circuitry, is to program the carrier detect output
using a resistor from the meter drive pin to the Vcc sup¬
ply. The carrier detect pin is an open collector output so
a pull-up resistor is required. The carrier detect is active
low, meaning that an RF input above the programmed
trip level will yield a low output {<0.1 V) at the carrier
detect pin. When the RF input is below the trip level (or
is detuned) the carrier detect pin will be at the supply
voltage. The trip level is set by the resistor value used
between the meter drive pin and supply. A resistor of
130 kfi sets the trip level to approximately -110 dBm at
the first mixer's input, which is roughly the 12 dB SINAD
point of the receivers with no external RF amplification.
It should be noted that the meter drive current will not
have the same linear 0.1 /iA/dB current-input level rela¬
tionship as when the meter drive is buffered as discussed
above, so an analog RSSI output is not really achievable
when Carrier Detect is used.
• Recommend MC33171 as the operational amplifier.
The MC33171 is a low-power single supply single op
amp with offset adjustment capability.
• V C q 1 = MC3363 supply (2 V to 7 V)
• v OUT = Vcc 1 + 'meter <Rf)
• Vqq 2 =* Op amp supply. Make this high enough to stay
within the op amp's common mode input range —
equal to Vcc** + 2.2 V for the MC33171. This voltage
also must be high enough to provide the maximum
VquT desired.
• R| s can be added to level shift the output, and is
optional. The output voltage will be adjusted down¬
ward by a factor of (Vcc 1 “ Vcc2)(Rf/R|s)-
• Compensation capacitor Cc is added to ensure stability
and will limit the circuit's response time.
• This circuit is not recommended for general purpose
AM detection.
Muting (MC3363 only)
Audio muting can be provided in two ways. The carrier
detect output can be DC coupled to the MC3363 muting
op amp input (Pin 15) and the op amp output can serve
to mute the audio. That is, the op amp output (Pin 19)
serves as a switch to ground in the audio signal path.
When the carrier level decreases below the carrier detect
trip point, the carrier detect pin will go to Vcc and
op amp output will go into saturation, muting the audio.
This yields a simple squelch with minimum external com¬
ponents and is shown in Figures 10 and 14.
Another way to mute the audio on MC3363 is to use
the op amp as an active filter for detecting noise above
the audio passband. The recovered audio is fed through
the active filter, rectified, integrated and compared to a
reference level. When the level rises above the reference,
a squelch gate is triggered. The data slicing comparator
on the MC3363 might be used as a squelch gate. This
noise triggered squelch would be executed similarly to
the squelch in MC3357/59/61 FM IF applications. (Seethe
MC3359 data sheet for details.) This type of squelch frees
the Meter Drive circuit to provide a linear output as noted
under "Meter Drive (RSSI)" above.
MOTOROLA
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AN980
Data Recovery
Both receivers contain a data slicing comparator which
provides data shaping and limiting of frequency-shift
keyed (FSK) serial data transmissions. The data slicer is
a non-inverting type, with the negative input terminal
biased internally to Vc<y2. Typically the data slicer is AC
coupled to the recovered audio pin via a 0.01 p F to 0.1 ftF
capacitor. Larger coupling capacitors can cause distortion
of the detected output and this is seen as negative slew
rate limiting in Figures 8 and 9. A pull down resistor from
the detector output pin to Vgg will reduce this effect if
objectionable. The comparator output is an open collec¬
tor so a pull-up resistor is required.
Comparator hysteresis is available by connecting the
comparator output and input using a high-valued resis¬
tor. This helps maintain data integrity as the recovered
audio becomes noisy, or for long bit strings of one polar¬
ity. Resistor values below 120 kiT are not recommended
as the comparator input signal will not be able to over¬
come the large hysteresis induced. Figure 8A shows data
jitter resulting from noisy demodulated data signal. The
improvement seen when hysteresis was added is shown
in Figure 8B.
• ;- c= - s-KHi-jKajw
• f 3ev - 3
• AC : 066
• >c; - : '■>
• To:: — Ae-vp-ec 3 5 C-C- rr. 5
• 5c-:.~ ~s;e — ■ 3 . a.
• A? :o—"vs:e-e4i
5 0fm<
Figure 8A. Noisy Recovered Data Signal
Causes Data Jitter
■ f- £3 • 633 h<- sa.s
• ’o«. - i
• AC cc-; ,'ii: 06“ ur
• Vr; 5 v
• :-ace - Recovers AuC-; SC: -v s.v
• 5;r;3r- tra;e — D« O.isu' ■ ' I . c
SOfaC lv 500 a s
Figure 9A. FSK Data Recovery at 1200 Baud
The maximum usable FSK data rate for any narrow¬
band FM system is typically 1200 baud subject to IF and
quadrature bandwidth and adjacent channel spacing lim¬
itations. The approximate bandwidth required to gener¬
ate or receive a frequency modulated signal is:
BW * 2 (f mo d + fdev> kHz, where f moc j is the mod¬
ulating frequency and fg ev ' s the frequency deviation.
This is known as Carson's Rule and is fairly accurate.
Any modulating signal which exceeds the available IF
bandwidth will be attenuated and/or distorted. For proper
recovery of square waves including the leading and trail¬
ing edges approximately the 7th harmonic should be
present. For a 1200 baud (600 Hz) square wave with fdev
= 3 kHz, fmod = 4-2 kHz (7th harmonic of 600 Hz square
wave), the bandwidth needed is: BW = 2(4.2 + 3) kHz
= 14.4 kHz = ±7.2 kHz, which is acceptable in narrow-
band FM channels. Figures 9A and 9B show the effect of
trying to pass a 9600 baud modulated carrier through a
narrowband channel, with resulting degradation of
recovered data.
MOTOROLA
8
AN980
For narrowband RF modems where 300 baud is ade¬
quate, an audio frequency shift keyed (AFSK) approach
is recommended. In this application two audio tones (for
Logic "0"and Logic "1") are modulated onto an RF carrier
and transmitted to the receiver, which reproduces the
audio tone sequence. The audio tones can be generated
at the transmitter and decoded after the receiver by the
MC145442/3 single chip 300 baud modems.
BREAOBOARDING
Do not attempt to build a high frequency radio circuit
using a wire wrap or plug-in prototype board. While the
MC3362 and the MC3363 are "tame" as high gain receiv¬
ers go, high frequency layout techniques are critical to
obtaining optimal receiver performance. This means
(typically) a one- or two-sided copper clad board with
adequate ground plane connected to Vgg potential. It is
also important that all Vqc interconnections are made
using copper traces on the board. Do not use "free float¬
ing" point to point wiring for the Vqq interconnections!
In general, keep all lead lengths as short as possible, with
an emphasis on minimizing the highest frequency path-
lengths. Decoupling capacitors should be placed close to
the 1C. If these techniques are not followed then the
receiver sensitivity and noise quieting will suffer, and
oscillations can occur.
APPLICATIONS CIRCUITS
Single Channel VHF FM Narrowband Receiver
The first application shown is of a complete single
channel VHF receiver operating at 49.67 MHz. This appli¬
cation includes a suitable circuit for running the first local
oscillator under crystal control on a single channel, which
is particularly useful for dedicated remote control links
and low cost two-way radios through 75 MHz. The circuit
contains a simple carrier level based squelch circuit and
audio amplification.
The 49.67 MHz receiver frequency is within the 49 MHz
USA cordless telephone band. Radios built for this band
may qualify under FCC Code of Federal Regulations Title
47, Part 15, for use by unlicensed operators. It is impor¬
tant to know the federal regulations concerning a partic¬
ular frequency channel or band of channels before a
receiver circuit is design (seethe notes on FEDERAL REG¬
ULATIONS, RECOMMENDED STANDARDS above).
Figure 10 shows the complete receiver schematic. The
LC network shown is used to match the input impedance
AN980
MOTOROLA
9
of the RF amplifier to 50 fl at this frequency. The amplifier
collector load is a single resistor for simplicity and in
order to enhance stability. The method of Figure 2 was
used to develop the crystal controlled oscillator circuit at
38.97 MHz. The RC integrator rolls off the audio above 2
kHz in order to minimize unwanted noise output. This
enhances receiver sensitivity and provides proper audio
deemphasis. The receiver, without the audio amplifier,
has 6.2 mA current drain at Vcc =* 5 V for a total dissi¬
pation of 31 mW. Using a 455 kHz filter with a 6 dB
bandwidth of ±10 kHz the receiver has a 12 dB SINAD
point of 0.28 ^V, modulation acceptance of 10.4 kHz and
distortion below 1.2% with f mod = 1 kHz and modulation
deviation f dev = 3 kHz. The maximum (S + N)/N ratio
obtained is 60 dB.
The MC34119 audio amplifier adds 3 mA quiescent cur¬
rent drain at 5 V, can deliver 250 mW into an 8 fl speaker
and has differential outputs which eliminate the need for
the typical large audio coupling capacitor, it also has a
chip disable input which provides muting and power
conservation.
Ten Channel Frequency Synthesized
Cordless Telephone Receiver
A demonstration receiver circuit has been built featur¬
ing the MC3362 and the MCI45160 dual phase locked
loop (DPLL). This receiver features frequency synthesis
to cover the ten channels allocated in the USA for cord¬
less telephone (CT) receivers in the 46 MHz (handset) and
49 MHz (base station) frequency ranges. The MC14516X
series DPLl's feature two complete loops which control
both the transmitter output and receiver first LO
frequencies.
MOTOROLA
10
AN980
Figure 13. Simple Interface of MC3362/3
To DPLL MC14516X
Figure 11 shows the complete schematic diagram. A
simple RF transistor amplifier is included to overcome
antenna and RF preselector losses. The output of the VCO
buffer is amplified by an external transistor amplifier so
that the VCO signal strength is large enough to drive the
receiver input pin (Fin-R) of the DPLL properly. Gain of
the VCO is set at approximately 400 kHz using the LC
values shown. The SB (Pin 3) of the MC145160 is
grounded to disable the transmit loop to simplify devel¬
opment of the circuit and reduce power consumption.
The DC voltage at the varactor control input of the
MC3362 (Pin 23} is adjusted to Vqq/ 2. The system ref¬
erence frequency of 10.240 MHz is generated in the
MC3362 second LO and fed into the Osc-in (Pin 18} of the
MCI 45160.
With a supply voltage of Vqq = 3 V and modulating
signal fmod = 1 kHz, f^ ev = 3 kHz the receiver yields
an input sensitivity of 0.6 £tV for 20 dB noise quieting
and 0.2 /xV for 12 dB SINAD from a 50 Cl source. The
audio distortion is less than 3 percent. The minimum
noise floor is less than 80 /iV and the maximum (S + N)/
N ratio is 53 dB.
There is a simpler way to interface the MC3362/3 to the
MC145160 DPLL as shown in Figure 13. The VCO signal
(about 400 mVpp with Vcc = 3 V using a pull-down
resistor of 3 kft from the MC3362 Pin 20 to Vgp) is fed
directly into the Fin-R input (Pin 16) of the MC145160.
With this configuration, the noise floor is raised to 245 piV,
10 dB higher than the circuit of Figure 11.
256 Channel Frequency Synthesized
Two Meter Amateur Band Receiver
A more traditional PLL frequency synthesizer approach
is needed to provide frequency flexibility and to allow
the MC3362/3 receivers to operate in the VHF "high band"
(130 MHz to 172 MHz). A receiver is shown which covers
the entire Two Meter (referring to radio wavelength) ama¬
teur radio band from 144 MHz to 148 MHz in 256 channels
spaced at 20 kHz. The complete receiver and PLL fre¬
quency synthesizer are shown in Figures 14 and 15. The
receiver achieved the same specifications as the
49.67 MHz MC3363 receiver discussed above.
The MC3363 receiver was chosen because squelch and
good sensitivity with minimum component count were
desired. To obtain good operation of MC3363 VCO above
75 MHz, the first local oscillator must be running well. To
ensure this, the Vcc supply voltage is kept above 3 V
which increases the current in the local oscillator circui¬
try. Extra current is also injected into the local oscillator
via pull-up resistors of 10 IcQ from each of the local oscil¬
lator tank pins to the Vcc supply. With the components
of Figure 14, the receiver VCO had an average gain of
1.5 MHz/V.
The VCO output is amplified and fed into an MCI2017
dual modulus prescaler which drives the input of the PLL
frequency synthesizer. The MC145152-1 PLL frequency
synthesizer was chosen for its ease of use and parallel
input format. The MC33171 bipolar operational amplifier
was chosen as the active integrator (loop filter) because
of its low power drain, offset adjustment capability and
ability to operate from a single supply voltage. The design
equations and assumptions used to determine loop filter
components are shown below. The MC145152-1 data
sheet and other sources go into much more detail on PLL
theory and performance.
Calculations of Loop Filter For VCO PLL
Frequency Synthesis
Assumptions:
f Q = 135.3 MHz (local oscillator center frequency)
f s = 20 kHz (channel spacing)
fb = 0.01 f s (loop bandwidth)
f rc = 20 fb (filter cutoff frequency)
§ = 0.707 (loop damping factor)
Vqd = 5 V (PLL supply voltage)
KVCO = 9 4 x rad/V (VCO gain, measured on
MC3363 receiver)
Cl = 0.1 fiF (active integrator component)
AN980
MOTOROLA
11
Results:
fb = 0.01 f s - 0.01 (20 kHz) = 200 Hz
frc - 20 f b « 20 (200) = 4 kHz
Kp * Vqd / 2w = 0.796 (phase detector gain)
= (^)" 610rad/sec
Nt = fo/fs * 135.3 MHz/20 kHz = 6765
Rl * <0 KvcO 7 (Cl w n 2 Nt> = 29.7 kfi « 30 kH
R 2 = 2 § = (w n Cl) - 23.2 kH ~ 24 kO
Cc - 4 = (2 R-|f rc ) - 0.017 fiF
With an 8 bit parallel input format several possible
switch settings and resultant countervalues and receiver
frequencies are shown in Table 2 below (Note: N t = NP
+ A, where P = 64 for the MCI 2017).
_ w n = 2?rfh
{ 2 S Z + 1 + [( 2 §Z + 1)2 + 1 ]C
Table 2 . PU. Frequency Synthesizer Switch
Settings and Frequencies
Switches
N
P
A
*
fvco
(MHz)
.zJMs
frx (MHz)
- *vco +
10.7 MHz
00000000
104
64
0
6656
133.12
143.82
00000001
104
64
1
6657
133.32
143.84
01000000
105
64
0
6720
134.40
145.10
01111111
105
64
63
6783
135.66
146.36
10000000
106
64
0
6784
135.68
146.38
10001101
106
64
13
6797
135.94
146.64
10011100
106
64
28
6812
136.24
146.94
11010001
107
64
17
6865
137.30
148.00
11111111
107
64
63
6911
138.22
148.92
Single Chip Weatherband Receiver
An application of the MC3362 as a simple receiver
tuned to the NOAA Weatherband (162.4 MHz to
162.55 MHz) is shown in Figure 16. The RF input is
applied directly to the mixer input, using a simple "L
network" to provide impedance matching of the mixer
input to 50 Q. The system sensitivity for 12 dB SI NAD
is 0.67 /iV at the input from a 50 n source in this appli¬
cation, which is as good as most inexpensive weather
cubes and the dual conversion design allows for excel¬
lent image protection to be provided.
AN980
MOTOROLA
12
0.01 fj. 1
Vdo - o^H
5 V, REGULATED
MCI 45152-1
Fin
LD
Vss
0SC jn
V DD
OSCout
RAO
A4
RA1
A3
RA2
A0
pR
A2
pv
A1
MC
N9
A5
N8
NO
N7
N1
N6
N2
N5
N3
IW I
£10.24 *2Qp
i^T mhz %
^x^/20 p
MCI 2017
|MC in V CC
'Sf'2?
OUT SIG.
V CC = 5V.
-J-O REGULATED
j; o.i
sj: 0.001 n
:»P R
330 0
ROM VCO
OUTPUT
(SEE FIGURE 14)
1. The values of R1, R2, Cl and Cq are very important in determining
the loop characteristics. Those value* calculated in the text gave
fairly "clean’’ VCO Control Voltage and VCO Buffered Output
signals.
2. Switches S1-S8 can be mini-DIP or hexadecimalty coded
thumbwheel switches. SI is the MSB. SB is the LSB.
3. Vcc3 «= 7.2 V minimum, to ensure that the MC3317Vs input
common mode range is not exceeded. An acceptable way of
configuring the entire receiver's power is:
VCC*. * V CC 3
R2_.XI
1 MC33171 g
HjlNULL NCp
T
RV2 ± Rt,7
VCO TUNING VOLTAGE
(SEE FIGURE 14)
Figure 15. 256 Channel VCO Control Using PLL Frequency Synthesizer
(162.40-16155 MHz, 50 H)
50 p
MC3362
U.LM
_eHIH
10.245 MHz 50 p=
—101-
mu Rata
SFE10.7MS2-A
SPEAKER
8 ft-32 fl
; TOKO RMC I
J 2A6597_ j
I Lp = 680,1.
! C p * IK) p
NOTES:
1. Make Vqq connections along a substantial portion of copper plane.
Do NOT use point to point wiring for V^c interconnections!
2. Vqc (MC3362) * 2 V to 7 V, regulated. First local oscillator wilt drift
if this supply is not regulated!
3. Vqq 2 (MC34119) *> 2 V to 12 V, must be well decoupled from the
Vcc source for the MC3363.
4. LI = 3 turns #18 AWG, .2* dia., .05' spacing, air wound.
5. Pins 1 and 24 are differential RF input, and are unmatched and used
singie-ended in this circuit. If single-ended input is used, be sure to
bypass the unused pin.
6. All capacitors in microfarads, inductors in Henries and resistors in
Ohms, unless otherwise specified.
Figure 16. MC3362 Application as a Tunable Weather Band Receiver
MOTOROLA
13
The first local oscillator is free-running in this appli¬
cation and the receiver is manually tunable over a range
of ±1 MHz. The oscillator's frequency and tuning range
are determined by the external tank circuit values chosen.
Keep in mind that the internal varactor diodes add
10-25 pF of capacity across the tank pins, depending on
the varactor control voltage applied.
This circuit is easily built to vefify receiver character¬
istics on the lab bench, but as shown is not suited for
mass production. The local oscillator temperature sta¬
bility is not nearly adequate in this free-running config¬
uration and microphonic pickup is difficult to avoid.
Before a narrowband receiver is production-ready, the
first local oscillator must be stable to within approxi¬
mately ± 100 Hz. The "First Mixer and Oscillator” section
provides notes on driving the first mixer using an external
oscillator signal above 50 MHz. The MC2833 FM trans¬
mitter 1C might serve as the local oscillator source up to
200 MHz.
SUMMARY
The high degree of integration and MOSAIC process
used in the MC3362/3 receivers give the radio designer
new levels of space and power economy, while providing
high performance and considerable design flexibility. The
receivers shown and alternate configurations discussed
should interest designers of cordless phones, VHF two
way radios, remote control receivers, wireless data links
and home security systems.
APPENDIX — DIRECTORY OF COMPONENT MANUFACTURERS
muRata-Erie
2200 Lake Park Drive
Smyrna, GA 30080
Toko America Inc.
1250 Feehanville Drive
Mount Prospect, IL 60056
Distributor — Digikey
Distributor — Inductor Supply
Coilcraft
1102 Silver Lake Road
Cary, IL 60013
California Crystal Laboratories
Comtec
Fox Electronics
International Crystals
Standard Crystal Corporation
(404) 436-1300
ceramic filters, coils
(312) 297-0070
quadrature coils, crystal filters,
coils, transformers
(800) 344-4539
(800) 854-1881
(800) 472-8421 (California)
(312) 639-6400
coils
(800) 333-9825
crystals
(602) 526-4123
crystals
(813) 693-0099
crystals
(405) 236-3741
crystals
(818) 443-2121
crystals
Motorola does not endorse the vendors listed.
This is a partial vendor list and no liability is assumed
for omissions or errors in address, product line or other information.
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me suit^rtrty of its products tor any particular purpose, nor does Motorola assume any liability arising out of the application or use of any pnxkjct or circuit,
artospedfically declaims anyarxl all liability, inchjdmg without fimitation consequential or incidental damages. ^ypcaT parameters canard do vary in cfiffwwt
applications. All operatng parameters, inchxSnfl Typical must be validated for each customer appficabon by customer's technical experts. Motorola does
not convevaiy license ixider its patent rights northe rights of others. Motorola products are not designed, intended, or authorized tor use as components in
systems intended tor sugical implant into the body, or other applications intended to support or sustain life, or lor any other application in which the failure of
the Motorola product could create a situation where personal njury or death may occur. Should Buyer purchase or use Motorola p roducts for any such
unintended or urauthorized application. Buyer shaft indemnify and hold Motorola and its officers, employees. subsieSanes, affiates, and cfistrtxrtors harmless
against al claims, costs, damages, and expenses, and reasonable attorney tees arising out of, dfreetty or intSrecJJy, any claim of personal injury or death
yiri >iin«TTtoort^fv-iinaiittv^ad usa. aver rf such daim alleges that Motorola was reqtioent rapardirq the design or maru^acture of the part.
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