Document text
FM Terminal Transmitter and Receiver
For the TH Radio System
By E. W. HOUGHTON and R. W. HATCH
(Manuscript received June 12, 1901)
The FM terminals form an important subsystein of TH radio, as the link
between the 0-10 mc baseband signal and the V+.l-mc FM signal. Severe
requirements arise from the design objective of 16 terminal pairs in tandem
in 4000 miles. The FM transmitter design uses a 6-kmc reflex klystron as
a frequency modulator, the output of which is heterodyned down to 74 mc by
another 6-kmc source. Automatic frequency control, with a 74-lS-mc crystal
oscillator as reference, provides the required frequency stability. In the FM
receiver, an IF amplijier-limiter is followed by a balanced FM discriminator
which uses parallel resonant discriminator networks. Improved linearity is
obtained by special design of a common interstage network. The video am-
plifiers are balanced and use high-performance electron lubes. The over-all
gain of a terminal pair is 8 db, between 124-ohm balanced video circuits.
I. INTRODUCTION
The relation of the FM terminals to the over-all TH system is de-
scribed briefly in a previous paper. 1 There are two basic types of termi-
nals: an FM transmitting terminal, which converts the baseband signal
into a frequency modulated signal centered at 74.1 mc; and an FM re-
ceiving terminal, which recovers the baseband signal from the FM signal.
The types of baseband signals to be transmitted 1 are shown in Fig. 1.
In each terminal appropriate amplification is provided at both the inter-
mediate and baseband frequencies to permit interconnection with other
parts of the Til system. FM terminals are required at the ends of a TH
route and at intermediate points where the baseband signal, or some por-
tion of it, must be added or dropped. The design is based on a maximum
of l(i terminal pairs in tandem in 4000 miles.
1587
1588 THE BELL SYSTEM TECHNICAL JOURNAL, NOVEMBER 1961
36112
(60 CHANNELS)
\\ \ MASTERGROUP 1 ~| | MASTERGROUP 2 1 1 MASTERGROUP 3
ALL TELEPHONE (i860 CHANNELS)
TELEVISION SIGNAL
NTSC TELEVISION
COLOR CARRIER
3.58 MC
| TELEVISION SIGNAL (WIDEBAND)
WIDEBAND TELEVISION
In [mi hinl Iii i iIiiiiIiiiiiii' il iiiiIii i ilii ii lrili i i n il m il l li l l i l ll i n i l iiill
01234567 89 10
BASEBAND FREQUENCY IN MEGACYCLES
Fig. 1 — The three types of baseband signals transmitted in the TH system.
II. DESIGN OBJECTIVES AND CONSIDERATIONS
2. l Over-all Telephone and Television Objectives
The over-all telephone (TP) and television (TV) objectives for 4000
miles of TH are based on extensive experience with previous systems.
From these over-all objectives, allocations were made early in the TH
development to the various portions of the system based on estimates
of the relative difficulty and expense which would be required to achieve
them. The allocations to the FM terminals arc summarized in Table I.
Table I — Objectives for 1G Terminal Pairs in Tandem
Telephone (TP)
Total noise (0 db TL) 31 dba
Fluctuation noise 28 dba
Cross modulation noise 28 dba
Television (TV)
Weighted signal-to-noise ratio* 62 db
Differential phase ±1 .25 degrees
Differential gain ±0.(i db
* The weighted signal-to-noise ratio is defined as the ratio of the peak-to-peak
signal to the weighted mis noise, where the weighting is a function of noise fre-
quency. For a detailed discussion see Ref. 2.
TERMINAL TRANSMITTER AND RECEIVER 1580
2.2 Signal Characteristics
Objectives such as those given in Table I can often be expressed in
alternative ways. For example, the linearity objective necessary to re-
strict cross modulation noise is stated above in terms of the resulting
noise in dba. For laboratory work, however, it is useful to state the
linearity objective in terms of the harmonic performance when a sine
wave of a given amplitude is applied. Conversions such as this depend
upon certain system parameters. Assumptions, subject to some change
when the over-all performance of the system could be determined, had
to be made regarding signal characteristics such as the peak frequency
deviation and the amount of pre-emphasis. The values of these quanti-
ties are shown in Table II.
2.3 Objectives for an FM Terminal Pair
Objectives for a single terminal pair were obtained by reducing the
total allocation of Table I by a factor of either 4 (12 db) or 16 (24 db) ;
the choice depends on whether a particular impairment could be ex-
pected to add on a random or on a systematic basis.
The principal design objectives for a single terminal pair are shown in
Table III. In the sections which follow some of them will be discussed
briefly.
2.4 Baseband Transmission
Essentially flat transmission is desired over the frequency band of all
three types of signal. The upper frequency limit is approximately 10 mc
for two; the lower frequency limit is set by the 60-cps component of a
television signal. To provide adequate phase linearity for the low-fre-
quency TV components, it is necessary to keep all low-frequency cutoffs
considerably below 00 cps. Based on experience with other video sys-
tems, a Low-frequency objective is expressed in terms of the distortion to
a 60-cps square wave. A distortion not exceeding 2 per cent of the nom-
inally flat top of the square wave is considered acceptable for a terminal
Table II — Signal Characteristics
Telephone signal
Peak frequency deviation 4 mc
mis frequency deviation 0.7 mc
Pre-emphasis 7.5 db
Television only
Peak frequency deviation 4 mc
Pre-emphasis (tentative) 12 db
1500 THE BELL SYSTEM TECHNICAL JOURNAL, NOVEMBER 19G1
Table III — Design Objectives for One FM Terminal Pair
Baseband transmission
Bandwidth
2 cps to 10 un-
Gain stability
it). 25 db
Peak frequency deviation
4 mc
Center frequency of FM transmitter
74.13 ± 0.1 mc
Harmonic performance*
Peak, deviation for applied sine wave
4 mc
Second harmonic with respect to fundamental
-49 db
Third harmonic with respect to fundamental
-51 db
Differential gain*
See Note
Differential phase
0.3 degree
Fluctuation noise
Message (at db TL)
16 dba
Television (weighted signal-to-noise ratio)
74 db
* Differential gain and harmonic performance are analytically related as shown
in Appendix A. The design objectives above for harmonic performance are set by
cross modulation requirements for telephone; they also insure adequate differ-
ential gain performance for television.
pair. To meet this, the low-frequency cutoff (3-db point) actually occurs
at about 2 cps. Transmission flatness of about ±0.1 db is the objective
for the band from 60 cps to 10 mc.
The objective of ±0.25 db for gain stability comes from two main con-
siderations. First, there is the desire to control net loss in toll telephone
channels within rather close limits. Second, all operating terminals at a
given point must have almost the same net loss as standby terminals.
Otherwise, switching from a regular terminal to a protection terminal will
cause hits in data signals.
2.5 Harmonic Performance
Nonlinearity in the terminals causes cross modulation in the TP sig-
nal, and differential phase and gain in the TV signal. It was found that
the linearity required for the telephone signal was controlling by a slight
margin. This led to the objective on harmonic performance given in Ta-
ble III.
2.6 Differential Gain and Phase
The NTSC color television signal uses a modulated 3.58 mc carrier to
transmit color information. Intermodulation between the low-frequency
luminance information in the signal and the color carrier causes ampli-
tude and phase variations in color carrier which are a function of the
luminance signal. These variations show up as distortion in the satura-
tion and chroma of the reproduced TV picture.
TERMINAL TRANSMITTER AND RECEIVER 1591
A special test signal 3 consisting of a low-level 3.58-mc tone and a
higher-level 15.75-kc tone is used to simulate the TV signal and to test
for distortion of this type. Variations in the amplitude and phase of the
3.58-mc tone as a function of the 15.75-kc tone are referred to as differen-
tial gain and phase. For tests of this type the peak-to-peak amplitude
of the low-frequency tone is normally made equal to the peak-to-peak
amplitude of the TV signal it simulates. It is then possible to specify
quantitatively the amount of differential gain and phase which corre-
sponds to a tolerable amount of color distortion in the TV picture.
Since differential gain and phase distortion occurs as the result of in-
termodulation, the amount of distortion which occurs in a particular
nonlinear system depends on the amplitude of the applied signal. Differ-
ential phase and gain distortion can be reduced by reducing the signal
amplitude, but at the expense of a poorer signal-to-noise ratio. A com-
promise between these two types of signal degradation is often possible
by reducing the amplitude of the low-frequency components of the TV
signal before transmission by means of a pre-emphasis network. An in-
verse network is used to compensate at the receiving end. Pre-emphasis
of the amount shown in Table II is proposed for this system. With this
pre-emphasis, the objectives for harmonic; performance shown in Table
III ensure that the over-all differential gain objectives will be met.
III. FM TRANSMITTER
The FM transmitter provides a +ll-dbm output signal, centered at
74.1 mc, which is frequency modulated proportional to the input base-
band signal. One volt peak-to-peak signal produces 8-mc pcak-to-pcak
frequency deviation.
To preclude excessive intermodulation, very little nonlinearity is per-
mitted in the frequency deviation vs voltage characteristic. This objec-
tive had a very strong influence on the selection of a modulation method.
Other objectives that had important influences on detailed design ap-
proaches were: the wide baseband, a conversion gain stability of ±0.15
db, and a carrier frequency (74.13 mc) stability of ±0.1 mc.
After a considerable exploratory development period during which sev-
eral more compact circuits were rejected because of marginal linearity or
bandwidth, a reflex klystron was selected as the frequency modulator.
A similar circuit using klystrons is successfully employed in the FM
modulator associated with the TD-2 radio system. 4 However, improve-
ments in the klystron and its associated circuits were essential to meet
the more stringent requirements of the TH system.
1592 THE BELL SYSTEM TECHNICAL JOURNAL, NOVEMBER 19G1
3.1 General Description
Selection of a klystron modulator essentially establishes the principal
accessory units. These are shown on Fig. 2.
The baseband signal, amplified by the video amplifier and applied to
the repeller of the deviated (DO) klystron, causes its frequency to vary
around a rest value of approximately 6174 mc. The deviated signal is ap-
plied through an isolator to the converter, where it is mixed with a 0100-
mc signal from the beating (BO) klystron. The output, a frequency
modulated wave centered on 74.1 mc, passes through a delay equalizer
(which compensates for the delay distortion of a tandcmly connected FM
transmitter and FM receiver), to an IF amplifier. The amplifier output
connects via coaxial cable to the other parts of the TH system.
To provide an input for the automatic frequency control (AFC) cir-
cuit, a small fraction of the output from the FM transmitter is abstracted
and amplified. Alternating samples of this FM signal and the output of
a crystal-controlled 74.1-mc oscillator are applied to the limiter-discrim-
inator. The output is a square wave with an amplitude proportional to
the frequency difference. This error signal is amplified and rectified in
the synchronous detector. The resultant voltage is applied to the BO
klystron with the proper polarity to reduce the average frequency error.
A photograph of the FM transmitter is shown in Fig. 3. The klystrons,
converter and other microwave devices are mounted in the FM gen-
erator panel. The video amplifier is at the bottom and the transmitting
IF amplifier is just above the frequency comparator panel. A further
description of the equipment features is given in a companion paper. 6
TRANSMITTING
IF
AMPLIFIER
IF OUTPUT
+ 11 DBM
TO ALARM
AND
PROTECTION
SWITCHING
Fig. 2 — Block diagram of FM transmitter.
TERMINAL TRANSMITTER AND RECEIVER
1598
IF AMPLIFIER
REFERENCE
OSCILLATOR
LIMITER
DISCRIMINATOR
IF AMPLIFIER
FREQUENCY
COMPARATOR
FM GENERATOR
VIDEO AMPLIFIER
AND JACK FIELD
Fig. :i — The FM terminal transmitter.
3.2 Video Amplifier*
The gain of the video amplifier is .sueh that a signal of 1 volt peak-to-
peak applied to its 124-ohin balanced input is increased to 5.5 volts peak-
to-peak at the repeller electrode of the DO klystron, which is sufficient
to deviate the klystron 8 mc peak-to-peak. At this output voltage, a
typical video amplifier has second and third harmonics that are below
the fundamental by ()() db and 75 db respectively.
To realize this performance, two balanced electron-tube amplifier
* Important contributions were made to this and other sections dealing with
the video amplifiers by H. ('. Hey.
1594 THE BELL SYSTEM TECHNICAL JOURNAL, NOVEMBER 10G1
stages arc used in a circuit configuration similar to that of the video am-
plifier described in Section IV. Low second- and third-harmonic distor-
tion results from the use of relatively lightly driven, high-current elec-
tron tubes in the two stages. Western Electric 448A tetrodes and 437 A
triodes are used, respectively, in the input and output stages. Additional
suppression of the second harmonic is achieved through the balancing ac-
tion of push-pull stages, which have cathode feedback to reduce and
stabilize residual unbalances.
To keep the gain-frequency distortion less than ±0.1 db up to 10 mc,
design emphasis is placed on minimizing spurious interstage capacitance.
Interstage resistances are limited to values which give a 3-db gain reduc-
tion at 14 mc. Interstage compensation, part of which is individually
adjusted, is used to achieve flat gain. The use of large coupling capacitors,
along with some additional phase compensation, limits 60-eps square
wave distortion to less than 1 per cent.
:j.:j Microwave Circuit
The microwave circuit consists of the two klystrons, an isolator, an RF
attenuator and a converter. The converter employs a silicon diode in an
unbalanced microwave network with internal resistance padding to re-
duce changes in performance with different diode characteristics. The
output power from the klystron is sufficiently high to make unimportant
the added conversion loss due to the padding. Variations in the output
amplitude of the FM wave are reduced by the limiting action obtained
in the converter. This limiting action is achieved by making the power
from the frequency-deviated oscillator (DO) substantially higher than
that from the undeviated beating oscillator (BO). The RF attenuator
determines the level difference, since the two klystrons are of the same
type and generate approximately the same power output. Presentation
of a well-matched impedance to the DO klystron is required for modula-
tion linearity, and is obtained by use of the microwave ferrite isolator,
as shown in Fig. 2.
Radiated microwave interference to or from the klystrons is precluded
by enclosing them in a shielded compartment into which all leads are
brought through microwave filters.
3.4 Reflex Klystron Modulator
To meet the stringent linearity objectives for the TH system a new
klystron, coded as the Western Electric 450A, was designed. It has a
limited tuning range (6000 mc to 6200 mc), a low-Q resonator (loaded
TERMINAL TRANSMITTER AND RECEIVER 1595
Q tt 100) and requires resonator, repeller and heater potentials of ap-
proximately 450, — 100 and (5.3 volts, respectively.
The loaded Q of the klystron is important in determining both the
linearity and the fluctuation noise performance. As shown in Appendix
B, the principal distortion terms due to nonlinearity in the voltage-fre-
quency deviation characteristic are proportional to Q 2 . On the other
hand, fluctuation noise, due to shot noise in the electron stream, is pro-
portional to Q~ v - (Appendix D). Thus, the selection of too high a Q leads
to excessive intermodulation products, whereas too low a Q causes ex-
cessive fluctuation noise. A compromise value of 100 was selected to give
the best over-all performance. With this, the second and third harmonics
are respectively at least 54 db and 56 db below a fundamental which has
a peak-to-peak deviation of 8 me. The FM components of the fluctuation
noise, as measured by an FM receiver, are flat with frequency above
about 25 kc. Below 25 kc the noise power increases with an approximate
1// law. Above 25 kc the rms frequency deviation due to the noise in two
klystrons is approximately 0.6 cps in a one-cycle band.
The deviation sensitivity of the klystron is approximately inversely
proportional to loaded Q, and essentially independent of baseband fre-
quency up to 10 mc. The reasons for this arc demonstrated in Appen-
dix C.
3.5 IF Amplifier
The two identical IF amplifiers (transmitting and AFC) shown in Fig.
2 each have three tubes. Each has a minimum gain of 21 db, a maximum
output of + 1 1 dbm, and a bandwidth of 58 to 90 mc between 0.3-db
points. The bias on the intermediate tube, supplied from an external
source, can be varied for manual gain adjustment or for gating the am-
plifier on and off as required in the AFC circuit.
Input, output and intermediate stages in this amplifier are almost
identical in electrical design to corresponding stages in the main IF am-
plifier of the radio receiver, described in a companion paper. 6
3.6 AFC Circuit
The AFC circuit, shown schematically on Fig. 4, is designed to hold
the average frequency of the outgoing FM wave at 74.1 ± 0.1 mc. A
low-temperature-coefficient, crystal-controlled oscillator operating at
74. ISO mc provides the basic reference against which the average fre-
quency of the outgoing FM wave is compared. By alternately gating the
74.1-mc reference oscillator and the AFC IF amplifier on and off at a
1596 THE BELL SYSTEM TECHNICAL JOURNAL, NOVEMBER 1961
to BO
EPELLER
C
6CV. POWER LINE FM
SYNCHRONIZING SIGNAL
INPUT INPUT
1 1 AAA
1
i
IF
AMPLIFIER
30 ^
SQUARE WAVE
GENERATOR
1
1
1
MERCURY
RELAY
tr
j i_i
"
)
1X1
AFC
MONITORS
74.1 MC
OSCILLATOR
AMPLIFIER
J 1 -^
LIMITER
r
i
DISCRIMINATOR
Fig. 4 — Functional schematic of the AFC system.
30-cps rate, there appears at the IF limiter input during one ^-second
interval a signal from the 74.1-mc oscillator, and during the next ^-sec-
ond interval a sample of the outgoing FM signal. The limiter section
equalizes the signal amplitudes, and the frequency discriminator delivers
to the audio amplifier a 30-cps square wave whose peak-to-peak average
amplitude is proportional to the frequency difference between the refer-
ence oscillator and the average (carrier) frequency of the outgoing FM
wave. After amplification the 30-cps wave is rectified in a synchronous
detector (the mercury relay), the dc output of which is related in ampli-
tude and polarity to the frequency difference. After filtering, this dc is
applied to the repeller of the BO klystron with a polarity which causes
its frequency to change in a direction to reduce to a small value the differ-
ence (error) in frequencies of the 74.1 mc oscillator and the outgoing FM
wave.
Since the instantaneous frequency of the FM wave is continuously
changing, nonlinearity in the AFC discriminator will give an average
output not linearly related to the average frequency of the FM wave.
This discriminator has a first-order nonlinearity of 2 per cent or less be-
tween 70 and 78 mc. On the basis that the rms frequency deviation will
be less than ±1 mc, the average frequency shift error will be less than
5kc.
The use of ac amplification following the discriminator makes the cir-
cuit insensitive to small drifts in the center frequency (corresponding to
TERMINAL TRANSMITTER AND RECEIVER lf)'.)7
a dc output of zero) of the discriminator. Synchronous rectification of the
amplified ac output restores sense so that the resultant voltage is applied
to the BO repeller with the correct polarity. Since the AFC open-loop
gain is typically 40 dh, uncorrected frequency differences of as much as
10 mc are reduced by AFC action to 100 kc.
When the outgoing FM wave contains a television signal, large 60-cps
and harmonically related voltages are present in the discriminator output
during signal sampling intervals. A dc output related to the true average
values of the signal voltage must be obtained for accurate frequency
control. This is accomplished by using negative feedback to linearize the
audio amplifier and by using a mercury relay switch for rectification.
The switch has negligible storage reactance in its output load so that its
dc output is proportional to the average value of the input.
When the. frequency of the local TH ac power differs significantly from
that of the remote power line against which the television signal is syn-
chronized, successive signal samples begin (and end) at different phases
of the OO-eyele ac component in the TV signal. Consequent "flicker" in-
terference 7 on television signals caused by the apparent shifts in average
frequency at the beat frequency rate are reduced to tolerable values by
the RC filter following the rectifier.
Other beat frequency effects are kept small by synchronizing the 30-
cps gating and rectification functions with local power frequencies. The
RC filter on the rectifier output has a 3-db cutoff at 0.005 cps. It gives
60-db loss at f) cps, the most annoying beat frequency, and reduces the
40-db AFC loop gain to unity at 0.5 cps. Transient response is optimized
by virtue of the 90° phase asymptote. Video phase shift, and conse-
quently the GO-cps square wave response of the FM transmitter, is only
slightly affected by the AFC action, which has an effect approximately
equal to that of a low-frequency cutoff at 0.5 cps.
IV. FM RECEIVER
The FM receiver accepts the 74-mc FM signal and delivers a balanced
baseband output signal which is 8 db above 1 volt peak-to-peak in a
124-ohm circuit for 8-mc peak-to-peak frequency deviation.
Performance requirements on linearity, baseband transmission, stabil-
ity and noise are comparable to those already discussed for the FM trans-
mitter. After careful study, the design described in the following sections
was selected as the best compromise between over-all performance and
ease of maintenance and adjustment. In its main features the design is
similar to that which has been used in the TD-2 radio system 4 for a num-
ber of years. Modifications in the detailed circuitry, however, have
1598 THE BELL SYSTEM TECHNICAL JOURNAL, NOVEMBER 19G1
achieved a considerable improvement in bandwidth, linearity and circuit
stability.
4.1 General Description
A block diagram of the FM receiver is shown in Fig. ">. The FM input
is applied to the receiver by means of 75-ohm coaxial cable. Provision
is made for an IF amplifier, identical with the two in the FM transmitter,
in case additional IF gain is required. In any event, the FM signal at +1
dbm is applied to an amplifier-limiter to suppress any amplitude modu-
lation of the signal, which otherwise would cause unwanted distortion
in the discriminator. In addition, the limiter action tends to maintain a
constant input power for the discriminator circuit. Without it, the dis-
criminator output would vary linearly with changes in input carrier
power to the FM receiver. This would cause undesirable variations in
the net gain of a terminal pair. The amplifier-limiter is electrically identi-
cal to the one in the broadband radio transmitter. 6 The mechanical de-
sign, however, is somewhat different, 5 to be in keeping with the plug-in
styling of the other FM terminal circuits.
The adjustable attenuator ahead of the discriminator is used to set
the sensitivity of the discriminator section to a standard value. The dis-
criminator recovers the baseband signal from the FM wave, in two steps.
First, networks which introduce amplitude slope across the IF band
produce amplitude modulation, which is proportional to the frequency
modulation of the input signal. Amplitude detectors then recover the
baseband signal, which is subsequently amplified in the baseband am-
plifier.
A photograph of an FM receiver is shown in Fig. 0, in which the in-
dividual units are easily identified.
+ 9 DBM
74 MC
FREQUENCY
MODULATED
SIGNAL
FROM
RADIO
EQUIPMENT *\
--*«-
-^
AMPLIFIER-
LIMITER
ATTEN-
UATOR
0-15 DB
S BASEBAND
SIGNAL
DISCRIM-
INATOR
VIDEO
AMPLIFIER
+ 8DBV
(1240. "
BAL)
IF
AMPLIFIER, _^
MAXIMUM
GAIN 22 DB
ADJUSTED FOR '
SENSITIVITY OUT
OF DISCRIMINATOR
OF 1V/8MC
DEVIATION
30CPS-10MC
TO
TELEPHONE
OR
TELEVISION
EQUIPMENT
VIDEO
;AMPLIFIER
MONITOR
-o ALARM
PROTECTION
SWITCH
(IF REQUIRED)
Fig. 5 — Block diagram of FM receiver.
TERMINAL TRANSMITTER AND RECEIVER
1599
Fig. (i — The FM terminal receiver.
4.2 FM Discriminator and Video Amplifier
Simplified schematics of the FM discriminator and of the video ampli-
fier arc shown in Figs. 7 and 8. The input signal is transformer-coupled
to two 448A electron tubes, vi and V2, which are operated in parallel.
Their combined output is developed across a common interstage, con-
sisting essentially of a parallel resonant circuit, and applied to the grids
of the 418A electron tubes, V3 and V4, which are driven in parallel. All
four tubes are stabilized by dc feedback and cathode compensation net-
works (zi — Z4) as described for the IF amplifier in the broadband radio
receiver. 6 The common interstage performs an important function in the
over-all design which will be described later. A 75-ohm test jack is pro-
vided to aid in interstage adjustment.
1600 THE BELL SYSTEM TECHNICAL JOURNAL, NOVEMBER 1961
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Tubes V3 and V4 are used as constant current generators to drive
separate parallel-resonant discriminator networks. The high-frequency
network is peaked at approximately 100 mc, and the low-frequency net-
work is peaked at approximately 50 mc. Each provides a large amplitude
slope across a wide band centered at 74. IS mc. Tuned at these frequen-
cies, the slope of one is positive and the other negative across the IF
band. The output voltages developed across the networks are amplitude
modulated with 180° phase difference between them. The diode de-
tectors, v5 and ve, recover the baseband signal from the amplitude
modulation, thus providing a balanced input to the balanced video am-
plifier which follows.
The video amplifier section (Fig. 8) consists of three balanced stages
in tandem. The first is a cathode follower stage employing 417A triodes.
The cathode follower is used to minimize the capacitance facing the
diode detectors. Parallel resonant traps are provided in the cathode cir-
cuits to reduce the amount of 74-mc carrier entering the amplifier stages
which follow. The second stage, with 448A tetrodes, provides the re-
quired voltage amplification and feeds the balanced output stage. Fixed
low-frequency compensation and adjustable high-frequency compensa-
tion are provided in the interstage. The output stage, using 418A tet-
rodes, is connected as a modified cathode follower circuit. The output
is increased by connecting both cathodes and plates to the load in an
arrangement which is used in the A2A television transmission system. 8
4.3 Design Considerations
To meet the over-all system objectives, the discriminator has to be
very linear, yet give an output large enough to provide an adequate sig-
nal-to-noise ratio at the input to the video amplifier. Furthermore, the
circuit must be readily adjustable to allow for manufacturing variations
and subsequently be stable with time. The manner in which the design
has been affected by these considerations is discussed in the following
paragraphs, in which the usual order is reversed by working from the
output toward the input.
A study of the noise and microphonics expected in the first stages of
the video amplifier leads to the establishment of an objective of 1 volt
peak-to-peak at the detector output, for a peak-to-peak frequency de-
viation of 8 mc. This level is sufficient to keep the fluctuation noise
contribution of the FM receiver at least 10 db below that of the FM
transmitter and at the same time to prevent microphonics in the first
stages of the video amplifier from degrading the television signal.
The use of 6AL5 diodes for the AM detectors provides a compromise
TERMINAL TRANSMITTER AXD RECEIVER 1603
among the following objectives: good linearity, high detection efficiency
and ease of replacement. The operation is between that of an averaging
detector and a peak detector. The over-all detection efficiency is approxi-
mately 40 per cent, close to that of an ideal averaging detector. Thus,
the capacitance in the output circuit for the diodes provides some peak-
ing action to compensate for the loss due to the forward resistance of
the diode. However, this capacitance must be kept low to minimize
video roll-off at 10 mc.
from the desired output of 1 volt peak-to-peak and the diode effi-
ciency of 40 per cent, the necessary change in IF signal amplitude as it
is tuned across an 8-mc band is about 1.25 volts for each discriminator
network. This change in signal amplitude is a function of the discrim-
inator networks and the signal currents provided by the preceding tubes.
Restrictions are imposed by the linearity objective and the interstage
capacitance which must be absorbed. This limits the maximum change
in impedance which can be achieved across the 8-mc band to about 50
ohms. Thus, peak signal currents of about 25 ma are required from the
driving tubes. Furthermore, this amplitude must be provided with low
harmonic content. The second harmonic, in particular, will be enhanced
with respect to the fundamental by the amplitude-frequency charac-
teristic of the high-frequency network. The detector output will there-
fore contain an error term due to the harmonics. Good harmonic per-
formance is required to permit accurate adjustment of the discriminator
with sweep signals, and to a somewhat lesser extent, to prevent distor-
tion to the normal signal.
The need for a large signal current with low harmonic content led to
the selection of the 418A tube for this application.
4.4 Discriminator Linearity
The discriminator networks have a substantial amount of curvature,
predominantly parabolic, as shown in Fig. 9. This curvature, if uncom-
pensated, would result in a nonlinear relationship between the incoming
frequency modulation and the resulting amplitude modulation. One
method of correction is to use more complex discriminator networks.
This was not selected because of the difficulty in controlling parasitic
capacitance and inductance. A second approach, used in the discrim-
inator for the TD-2 system, is to select network designs such that the
parabolic curvatures of the two sides are equal. The predominant second-
order modulation products then tend to cancel each other in the bal-
anced output from the detectors. The major difficulty with this approach
is the amount of balance required. An analysis of the networks in the
1604 THE BELL SYSTEM TECHNICAL JOURNAL, NOVEMBER 1901
3.0
UJ
N
J i.o
<
tr
§0-9
LOW- FREQUENCY
NETWORK
^-^. f n = 56 MC
(a)
Q=4.3
V A
HIGH-FREQUENCY
NETWORK v /
f = 94 MC \V
r-
/'
4
C
= 4.8 S\
~^V
4
/
'
x
^
t'
X
<
<^
S5r:= ^:
-— — "
*■*
-■^
z
:\
(b)
60 65 70 75 80 85 90
FREQUENCY IN MEGACYCLES PER SECOND
Fig. 9 — (a) Discriminator network characteristics; clashed curves show effect
of common interstage; (b) typical discriminator linearity.
TH discriminator indicated that the second-order modulation products
from one side of the discriminator would be about 30 db above the de-
sired objective. Although a 30-db balance is obtainable at the time of
adjustment, it is difficult to hold between maintenance intervals since
it depends on the relative gains of V3 and V4.
Another approach,* the one used in this design, also requires the net-
works to have equal parabolic curvatures. However, instead of depend-
ing on precise cancellation in the output, a compensating parabolic
shape of opposite sign is introduced in the common interstage ahead of
V3 and V4. In this way the linearity of each side of the discriminator is
substantially improved, as shown in Fig. 9. With this design the non-
linearity of each side exceeds the over-all objective by only about 10 db.
This reduces by a considerable amount the effect of gain changes in the
driving stages V3 and V4. For example, without the common interstage,
only 0.5 db of change in the relative gains after the initial balance would
* This approach was suggested by N. E. Chasek of the Radio Research De-
partment at Bell Telephone Laboratories.
TERMINAL TRANSMITTER AND RECEIVER 1605
cause the objective to be exceeded. With the common interstage, the
acceptable variation is 4 db. Thus, the common interstage substantially
increases the time stability of the second harmonic balance.
The procedure used in designing the discriminator networks is out-
lined in Appendix E. This design was modified slightly on the basis of
experimental results to obtain the typical linearity characteristic shown
in Fig. 9(b).
V. POWER SUPPLIES
Supplied with regulated, reliable 220-volt, 60-cycle ac inputs, the
power rectifiers give dc voltages of —11, +135 and 4-220 volts for the
KM receiver; -11, -170, +135, +220 and +450 volts for the FM
transmitter. Electronic regulation is counted on to reduce bobble and
consequent television flicker interference to acceptable levels. This
stability also keeps within limits, for extended time intervals, varia-
tions in FM deviation, FM sensitivity, and second harmonic balances.
Additional details of the power supplies arc given in lief. 9.
VI. MONITORS
To preclude disruption or excessive degradation in service due to
failure in an FM transmiter or an FM receiver, the failed unit is auto-
matically replaced by a standby unit. Status information for initiating
this protection switching action and for registering alarms is obtained
from monitors on space current in video amplifier tubes, on IF carrier
power at the FM transmitter output, on three significant parameters in
the AFC system, and on rectified carrier level in the FM receiver.
(}.i Video Amplifier Monitors
A considerable simplification in instrumentation of video amplifier
monitors is based on the fact that any reduction in gain, or increase in
harmonic distortion, will probably be accompanied by a change in space
current in one or more electron tubes. Cathode voltages (which are pro-
port ional to space currents) from all tubes are added, and the sum is
applied to the input of a differential dc amplifier. A change in this sum
resulting from a change in any cathode voltage of 30 per cent or greater
will initiate an alarm and a protection switching order. Even though the
response time of this monitor is around two milliseconds, it is not oper-
ated by television waveforms having frequency components as low as
(>0 cps because of the push-pull action in the balanced video amplifier.
KlOG THE BELL SYSTEM TECHNICAL JOURNAL, NOVEMBER 1961
6.2 IF Carrier Monitors
Failures in klystrons or the transmitting IF amplifier in the FM trans-
mitter are detected by a reduction in output from an IF carrier level
detector connected across the outgoing IF line.
By monitoring the sum of rectified voltages at the discriminator out-
put in the FM receiver, information is obtained on the status of IF
carrier input and of the electron tubes and circuits in the discriminator.
This voltage sum is combined with the sum of the FM receiver video
amplifier cathode voltages and applied to a video amplifier monitor cir-
cuit of identical design to that used in the FM transmitter.
6.3 AFC Monitors
Because of the large time constant in the output filter, a failure in
the AFC control circuit will not cause an immediate change in the aver-
age frequency of the outgoing FM wave. This allows the use of three
reliable, though relatively slow-acting, sensitive meter relays for mon-
itoring this circuit. These relays monitor: (a) the peak-to-peak frequency
error voltage, (b) the rectified carrier level, and (c) the 30-cps gating
voltage. Failures in the audio amplifier or synchronous rectifier will
cause monitor (a) to initiate alarm and protection switching orders
whenever the klystrons drift sufficiently to create a 1.5-mc shift in dif-
ference frequency. However, monitor (a) is a null indication and so will
not be activated by failures in the gating circuits or in the limiter-dis-
criminator and preceding IF circuits. Failures in these circuits arc de-
tected by monitors (b) and (c).
VII. TANDEM PERFORMANCE
The baseband-to-bascband performance of a typical FM transmitter
and FM receiver when connected in tandem is discussed in this section.
This performance, compared with values in Table III, will show how well
the design objectives have been met.
It should be noted that "typical" back-to-back performance means
the average performance that can be expected during the time intervals
between maintenance adjustments on individual FM transmitters and
receivers. These specifications draw from experimental data in attempt-
ing to give the most probable performance that can be expected over a
reasonable time interval after maintenace adjustments have been made.
They observe the practical restriction that terminals cannot be con-
nected back-to-back in the field to adjust paired terminals for optimum
performance.
TERMTXAL TRANSMITTER AXD RECEIVER 1007
7.1 Envelope Delay Distortion {EDD)
Measured as the change in phase shift of a 278-kc tone while the IF
center frequency is swept between 04 mc and 84 me, the IF delay dis-
tortion (excluding the delay equalizer shown in Fig. 2) has the typical
characteristic shown on Fig. 10(a). Odd-order EDD at 04 and 84 mc is
approximately —5.3 and +5.3 m/xs respectively, and even-order EDD
is approximately +8.7 m^us at both frequencies. To this total the klys-
tron and its microwave circuits contribute almost nothing, the FM
discriminator contributes about half, and the remaining IF circuits
(amplilicr-limiter and transmitting IF amplifier) contribute about half.
With the delay equalizer, the distortion is reduced by a factor of at
least five.
7.2 Harmonic Distortion
Typical curves of second and third harmonic performance (exclusive
of the delay equalizer) arc shown in Fig. 10(b). At low frequencies the
harmonic performance is flat with frequency, and is primarily due to
nonlinear voltage-frequency characteristics in the klystron and in the
discriminator. At higher frequencies, however, second and third har-
monics tend to increase in proportion to the frequencies where the har-
monics fall. This is primarily due to EDD in the IF circuits. This source
is reduced by delay equalization until it is negligible compared to the
low-frequency asymptote. The performance then becomes that shown
by the dash lines in Fig. 10(b).
The limiting low-frequency value of —70 db for the third harmonic-
to-fundamental ratio comes from systematic contributions by two video
amplifiers, the DO klystron and the FM discriminator; typical values
are respectively -0.0004 (-68 db), +0.0015 (-56 db) and -0.001
(—00 db). The second harmonic contribution from the same four units
depends upon the status of balance in each unit. The low-frequency
asymptote of —50 db was obtained by taking the root-sum-squares
(R.S.S.) of limiting design values of ±0.001 (-00 db), ±0.002 (-54
db) and ±0.002 ( — 54 db) in each of two video amplifiers, the DO
klystron and the FM discriminator, respectively.
7.3 Differential Gain and Phase
As discussed previously, the harmonic performance given above in-
sures adequate differential gain performance for television.
Without delay equalization, the differential phase characteristic has
the same broad structure shape as the EDD characteristic of Fig. 10(a),
-mo
<<0Z
><-
-0.5
0.'
64 69 74 79 84
IF FREQUENCY IN MEGACYCLES PER SECOND
0.01 0.05 0.1 0.5 1 5 10 50 100
HARMONIC FREQUENCY IN MEGACYCLES PER SECOND
0.05 0.1 0.5 1 5 10
VIDEO FREQUENCY IN MEGACYCLES PER SECOND
10 100
•CYCLES
10 100
KILOCYCLES
10 100
-MEGACYCLES
CENTER FREQUENCY OF NOISE BAND
Fig. 10 — Tandem characteristics for an unequalized pair: (a) IF delay dis-
tortion, (b) typical harmonic distortion for 8-mc peak-to-peak deviation, (c)
typical baseband response, (d) fluctuation noise.
1608
TERMINAL TRANSMITTER AND RECEIVER 1609
but with reversed .sign, since it is also the change in phase shift (in this
case at 3.58 mo) while the carrier is swept by a low-frequency (15-kc)
video voltage. Odd-order and even-order differential phase are each
about 2.5° at ±4-mc peak deviation. Since the low-frequency compo-
nents of the TV signal are less than this by 12 db due to pre-emphasis,
the odd- and even-order differential phase experienced by the TV signal
is reduced by factors of at least 4 and 16 respectively. Differential phase
is further reduced by the delay equalization, and hence is well below
the 0.3° objective.
7.4 Baseband Amplitude Response
Low-frequency baseband response, determined by the characteristics
of the two video amplifiers, is typically down 3 db at 2 cps. This, with
some added low-frequency phase equalization, gives a 60-cps square
wave response having low-frequency slope of less than 2 per cent. High-
frequency baseband response, typically ±0.15 db at 10 mc, is the result
of a number of significant contributions.
As described in previous sections, video response in the klystron and
its microwave circuits is substantially constant. Response in each video
amplifier is adjusted to be constant up to 10 mc within ±0.1 db.
The EDD of Fig. 10(a) produces a video roll-off of 0.06 db at 10 mc.
In contrast to harmonic distortion, amplitude response is made worse
by delay equalization when the distortion and the equalization are
separated by a limiter. Thus, equalization in the FM transmitter, which
introduces an equal and opposite delay distortion, adds another 0.06
db to the video roll-off at 10 mc.
Contributing directly to baseband gain-frequency variations, even-
order IF gain variations in a typical transmitting IF amplifier account
for ±0.03 db. Though somewhat more subtle to observe experimentally
because of limiting action, the same basic mechanism in the amplifier-
limiter contributes approximately the same video gain deviation. Lin-
earity and detector response in the FM receiver results in a systematic
even-order characteristic of about —3 per cent at 64 mc and 84 mc, con-
tributing — 0.3 db to the video response at 10 mc.
Summing up the systematic contributions and adding the random con-
tributions on an rss basis leads to a typical characteristic of —0.4 ±
0.15 db at 10 mc. The systematic characteristic is equalized to give the
nearly flat response shown on Fig. 10(c).
7.5 Low-Frequency Noise
Noise below 300 kc causing interference in TV signals comes from
mechanical vibration in electron tubes, from "1//" cathode emission
1610 THE BELL SYSTEM TECHNICAL JOURNAL, NOVEMBER 1961
fluctuation noise in klystrons, from power rectifier output voltages hav-
ing "hum" components which are multiples of power-line frequency,
and from "bobble" components which are random variations in power
rectifier dc outputs at rates of 1 to 30 cps.
Rugged mechanical design in the klystrons and maintenance of rela-
tively high signal levels at the FM receiver video amplifier input mini-
mize mechanical vibration noise. Considering frequency weighting, TV
circuit clamper characteristics, and 12-db pre-emphasis for the com-
bined service, the contribution by "1//" cathode emission noise over
the band of to 25 kc can be neglected since it is more than 6 db below
weighted rms fluctuation noise in the rest of the TV band.
From the standpoint of deriving objectives and evaluating perform-
ance, an especially troublesome problem centered around the inevitable
small, random variations in the60-cps ac power. This power line "bobble"
creates "flicker" interference in TV signals which is most annoying at
repetition rates around 5 cps. 7 Well-balanced video amplifier stages
minimize sensitivity to additive components; cathode feedback and
high-current operation of electron tubes minimize the nonlinear gen-
eration of bobble modulation components. However, to keep flicker
interference within acceptable limits, the electronically regulated power
supplies are counted on to reduce the effects of power line bobble by
50 db to 60 db. Electronic regulation plus dc operation of electron tube
heaters and adequate filtering of unregulated power supplies control
the amount of "hum" interference.
As a final result, the weighted interference to television by power line
and other low-frequency noise in a terminal pair is expected to be 80 db
below the peak-to-peak TV signal.
7.G Fluctuation Noise
FM noise originating in the FM transmitter predominates over all
other sources of fluctuation noise in FM terminals by at least 10 db. This
noise originates in the two klystrons where it is generated by random
(shot) processes in the electron beam.
Allowing for the frequency weighting of the interfering effect, for the
action of typical TV circuit clampers and for the 12-db pre-emphasis,
the klystrons introduce a fluctuation noise having the typical relative
interference effect shown on Fig. 10(d). The integrated value of rms
fluctuation noise in a terminal pair is 81 db below the peak-to-peak
TV signal amplitude (8 mc peak-to-peak).
The relation of FM terminal noise to the over-all system noise for the
telephone signal is discussed fully in Ref. 1. Briefly, noise interference
TERMINAL TRANSMITTER AND RECEIVER 1611
introduced by terminals is most serious in the telephone master-group
at the lower end of the baseband. In a 3-kc band the rms frequency de-
viation due to noise is typically 35 cycles. Since an rms frequency devia-
tion of 2.828 mc produces +8 dbm at the receiver output, the noise at,
this point is —90 dbm. At this point, the transmission level for the
lowest level mastergroup is —25.5 db. Therefore, the noise is —64.5
dbm at db TL; this is 17.5 dba in these "noisiest" telephone channels.
If these channels remained at the same low-level end of the band (no
frogging) for 1(5 terminals connected in tandem, the noise would add
randomly to give a net noise meter reading of 29.5 dba at db TL.
7.7 Gain Stability
Electronic stabilization of the most critical dc power supplies has re-
duced to insignificance the gain variations from these sources. However,
power line voltage variation of ±1 per cent (typical stability) will cause
±0.2-db variation in the sensitivity of an FM receiver due to unregu-
lated heater supplies. This random variation, added to systematic gain
variations (estimated below as less than 0.05 db in one month), supports
the expectation that gain changes of less than ±0.3 db will occur when
an FM transmitter or an FM receiver is replaced by a standby unit
through protection switching action in the field.
Presently available data on gain stability with time are meager, being
the result of experimental observations over several month intervals on
terminals in which the ages of all electron tubes were considerably under
their life expectancy. Nevertheless, the performance under these condi-
gave a reasonable degree of confidence in estimating that in a one month
interval a typical terminal pair will have a systematic gain variation of
— 0.05 db and random variations not exceeding ±0.2 db. Sixteen terminal
pairs in tandem would then have a net gain variation of —0.8 ± 0.8 db
in a one-month interval.
APPENDIX A
Linearity, Differential Gain, Harmonic Distortion
Analytical expressions relating harmonic distortion to linearity and
differential gain are obtained through the coefficients in the power series
expansion for an output quantity (voltage, frequency deviation, etc.)
as a function of input quantity. For example, consider the output vs.
input characteristic
Wo = Oi» + a-.i>' 2 + art'* + • • •, (1)
1612 THE BELL SYSTEM TECHNICAL JOURNAL, NOVEMBER 1961
where the input, u, consists of a steady -state shift (or dc component), 8,
added to a sinusoidal variation, E cos pt; i.e., v = 8 + E cos pt. Then,
omitting dc terms, the output will contain the following sinusoidal
components:
where
v = Fi cos pt + F 2 cos 2 pt + F 3 cos 3 pt +
Fi = E\a x + 2a 2 S + 3a 3 5 2 + • ■ •]
^3 =
(2)
-»R
? +
Differential gain is defined as the ratio of the small-signal gain (at
fundamental frequency) for any shift, 8, to the gain when 5 = 0. Dif-
ferential gain, as a function of 8, then has the form
Differential gain = 1 + L\8 + Lafi,
where, from the expression for F\ in equation (2),
(3)
L\ = — and
0.1
T 3o 3
Li-i = .
Differential gain is normally expressed in db as 20 log (1 + L\8 + L 2 8-).
Nonlinearity is closely related to differential gain and is defined as
First-order nonlinearity = L18 X 100 per cent
Second-order nonlinearity = L->8- X 100 per cent.
(4)
(5)
Harmonic performance can be expressed either in terms of the power
series coefficients in (1) or the linearity coefficient in (3) as shown below:
. 2
Second harmonic
infill + *S|-feY
|_2ax x 2ai \aj
Fundamental
The third term above is frequently negligible, in which case
F,
B*4
Third harmonic _ F 3 _ .- ,
Fundamental Fi
(G)
(7)
= E*
TERMINAL TRANSMITTER AND RECEIVER
1613
APPENDIX B
Klystron Quasi-Stationary Frequency Behavior
This appendix develops analytical relationships defining the FM
properties of the 450A reflex klystron.
b.i Linearity of Steady-State Frequency Shift
The detailed analysis of the behavior of reflex klystrons given in Ref.
10 suggests that a circuit analogue like that shown in Fig. 11 can be used
to explain steady-state frequency behavior.
The phase shift, 6, in the drift space is a function of repeller voltage,
/•, and oscillating frequency, a>, while the phase shift, <p, in the resonator
is a function of frequency only. To have sustained oscillations at any
-IllK
HI'
Vn
REPELLER
MODULATING VOLTAGE
ELECTRON
STREAM
RESONATOR
WAVEGUIDE LOAD
4 »V V
V(t)
DRIFT SPACE
y INPUT
DRIFT SPACE
OUTPUT
REPELLER
DRIFT SPACE
OUTPUT
v(t)
♦
A14PI ipipo 2 PHASE j
iiuiTBB ' — "°" IMODULATORl
LIMITER | a I
DRIFT SPACE
(icos [a>t + v(t) + 0(t)]
COUPLED I
LOAD >
AND--»->R
INTERNAL >
LOSSES \
RESONATOR
DRIFT SPACE
INPUT
Ecos[o; t + ^(t)]
-A3
lb)
Fig. 11 — Circuit diagrams of reflex klystron: (a) schematic, (b) feedback loop.
1614 THE BELL SYSTEM TECHNICAL JOURNAL, NOVEMBER 1961
frequency, the gain around the oscillator loop must be unity with a phase
shift of zero. The equivalent amplifier-limiter action of the drift space
takes care of satisfying the unity gain requirement. To satisfy the zero
phase shift requirement, the oscillation frequency stabilizes at a value
such that the resonator phase shift is equal and opposite to the drift
space phase; i.e., <p + 6 = 0. If a change is made in the repeller voltage
to v + di> such that the delay in the drift space increases and the drift
space phase changes to + dd, then the oscillating frequency will decrease
sufficiently to give an equal and opposite phase change in the resonator.
Nonlinearity in the frequency-voltage relationship results primarily
from the nonlinear relationship between frequency and phase in the
resonator and to a lesser extent because the changes in drift space delay
are not linearly related to changes in the repeller voltage. These rela-
tionships will be evident from solutions to the differential equation :
d<p + dO = (8)
from which
and
'£,/„ + *? ,/, + *? </„ = <) (9)
dd) ou ow
ae
dco dv
di< d^ dd_
do) dco
(10)
In Rcf. 10, p. 593, the relationship between drift space phase and
electrode voltages is given as
where the voltages V and V r are those shown in Fig. 11. Also, assuming
that the resonator behaves as a simple parallel resonant circuit,
— arc tan Q
[=
12
These relationships can be different iated and the resulting expressions
expanded in power series which are valid in the vicinity of the operating
point of the klystron. When the first few terms of these series are sub-
stituted in (10), an expression for the small-signal FM sensitivity as a
function of the frequency shift, 8 = u — co r , and repeller voltage,
TERMINAL TRANSMITTER AND RECEIVER 1615
V r , is obtained. The repeller voltage can be eliminated as a variable in
the expression by substituting V T = Y r n + aS, where a = dv/du evalu-
ated at co = co r and V r o is the repeller voltage corresponding to an oscil-
lating frequency co r . The first few terms then become
r/co_ co r i |~, f*Q ^ 1 \± + . 8 <?
(f+0
-41
- do C0 r 2 J ,
dl> (Vo+Vro){2Q , ,\[_ \»0 /<* 2Q + O «,*_!, (13]
where
5 = co — C0 r ,
co r = resonant frequency of cavity in radians/ sec,
do = drift space phase angle in radians, at co r ,
Q = cavity Q, and
T'o + Vra = dc potentials applied to klystron as shown in Fig. 11.
Comparing this expression with (3) we write
Differential sensitivity = 1 + Li<5 + L 2 8 2 (14)
where
u ■ -c« - i
and
u
8Q a
(2Q + e W
Typical :
numerical v
alucs f<
>r these
1 quantities are:
co r
= 2tt :
>< () X
10 9 ,
00
= m
cycles
= 17.3 radians,
Q
= 100,
and
V +
Vr
= 550:
i
which
give
t to
Differential sensitivity =» 1 - 22- + 37,000—.
co r co r
For a ±4-mc frequency deviation (8/u r = 4/6000), the first -order non-
1616 THE BELL SYSTEM TECHNICAL JOURNAL, NOVEMBER 1961
linearity is ±1.5 per cent, and the second-order nonlinearity is 1.6 per
cent. The first-order nonlinearity is primarily due to the nonlinear rela-
tionship between drift space phase shift and repeller voltage, while the
second-order nonlinearity is predominently caused by the nonlinear
phase shift in the resonator as a function of frequency.
b.2 Second Harmonic Balance
It is evident from the expression for F 2 /Fi in (6) that the second
harmonic can be balanced to zero by selection of the operating point 5,
such that 5 = 5 = -Li/2L 2 . In terms of (14) a second harmonic
balance can be obtained by means of a small change in the klystron
repeller voltage so that the quiescent oscillating frequency is slightly
different from the resonant frequency of the cavity, the amount of this
difference being given by
i-i
nu
co r 16Q a
= 3.0 X 10"
2Q + O
This corresponds to shifting to an operating point which is approxi-
mately 1.8 mc above the resonant frequency of the cavity.
b.3 Experimental Adjustment for Second Harmonic Balance
As can be checked by differentiating (13) and setting the resulting
expression equal to zero, minimum differential sensitivity occurs at the
same value of 5 as was found for zero second harmonic. This principle
is employed in a field adjustment procedure for "optimizing" klystron
linearity: repeller bias is adjusted to minimize the small-signal FM
deviation produced by a small voltage variation.
b.4 Second-Order Balance Stability
Asa Function of Repeller Voltage. It has been shown above that it is
possible to adjust the klystron bias so as to obtain a second harmonic
balance. For such an adjustment to be worthwhile, however, the operat-
ing point must stay within bounds. For example, the ratio of the second
harmonic to fundamental is given in (6) as
Fi
s(l*+|«),
TERMINAL TRANSMITTER AND RECEIVER 1617
where for E = 4 mc, the requirement on Fo/Fi is —54 db* or 0.002.
Substitution of the numerical values above gives a requirement on the
frequency shift of 1.8 ± 1.0 mc. With a repeller sensitivity of about 1.45
mc per volt, the voltage stability requirement is ±0.7 volt for the — 100-
volt repeller supply.
As a Function of the Resonator Voltage. The resonator has a sensitivity
of approximately 0.5 mc per volt. Therefore, the frequency stability
requirement of +1.0 mc, just derived, imposes a voltage requirement of
about ±2.0 volts on the 450- volt resonator supply. Thus, regulated
supplies are required for both the repeller and resonator to maintain the
required second-harmonic balance.
As a Function of Resonant Frequency. If temperature or tuning changes
the resonant frequency of the cavity, the phase shift in the cavity is a
function of two variables: the oscillating frequency, co u , and the resonant
frequency, o> r . With V constant, (9) becomes
dip , dip . . dd .
— f/o> r + — f/COu + j— du) ■=
OW r OOJu f/wo
where
dip dip
di>) r dcou
for o>o ~ o) r
Substituting, and solving for duo/du r gives
dtp
f/co» don.
(lu) r dtp dd
For the 450A klystron
dip/du « -^ « 5.3 X 10 9 second
o-v
de/du w — f*> 0.46 X 10~ 9 second.
Wo
Therefore,
^?«0.92
<IU) r
* Only a portion of the total requirement of —49 db is allocated to the klystron.
1018 THE BELL SYSTEM TECHNICAL JOURNAL, NOVEMBER 1961
and as the resonant frequency changes, the oscillating frequency tends
to follow so that the change in the desired 1.8-mc offset is only 8 per cent
of the change in the resonant frequency.
Thermal and mechanical stabilities in the 450A klystron are such that
second-order balance degradation due to rest frequency shifts caused by
resonant frequency changes are generally negligible compared to the
degradation due to resonator and repeller supply voltage changes.
B.5 Third -Harmonic Distortion Performance
The low-frequency third-harmonic performance of the 450A klystron
with a resonator Q = 100 and for a peak frequency deviation of 4 mc is
given by (7) with E = 4 mc and
8Q :t
Jj (2Q + 6 W
Thus,
Third harmonic _ .,2 SQ'
Fundamental l2w r 2 (2Q + 80)
= 0.00137 or -57.2 db.
This is db better than the objective of -51 db.
appendix c
Klystron Dynamic Behavior — Video Response
It was tacitly assumed in Appendix B that the steady-state frequency
deviation behavior is also applicable when the incremental repeller
voltage is a function of time. This assumption has been verified experi-
mentally; frequency deviation is substantially independent of video signal
frequency up to at least 10 mc. Similarly, harmonic distortion depends
upon Q as predicted by the steady-state analysis.
This dynamic behavior may also be verified theoretically. The essence
of this theoretical analysis will be outlined here to show how the inte-
grating action of the resonator dynamically converts phase modulation
to frequency modulation. Appendix D shows how this same action con-
verts noise in the electron stream to an FM deviation noise in the output.
When a signal voltage is applied to the repeller in Fig. 11(b), a phase
modulation of the carrier is produced in the closed loop. Finding a rela-
tionship between the phase modulation (output) and the modulating
signal (input) is analogous to the problem encountered in any closed-loop
TERMINAL TRANSMITTER AND RECEIVER 1(319
system such as a feedback amplifier. In principle, the loop between (1)
and (3) on Fig. 11(b) is temporarily opened, and an arbitrarily amplitude
and phase modulated signal,
e(t) = Ex(t) cos M + \K0], (15)
is introduced at 1. This signal is traced around the loop and appropri-
ately modified until the output at 3 is expressed in terms of the arbitrary
wave introduced at 1. At this point the input and output waves for the
open loop can be equated to determine the closed-loop performance of
the system. Since the output signal, as well as the input signal, will have
the form of an amplitude and phase modulated wave, the amplitude
modulation and phase modulation can be separately equated. Only the
results for the phase modulation terms are of interest here. Some simpli-
fication is achieved by working in the frequency domain. Thus, the fol-
lowing definitions are made.
»SV(w) = frequency spectrum of \p(t), and
Se(o)) = frequency spectrum of d(t), the phase modulation intro-
duced by the modulating signal, V(t).
The spectrum of the phase modulation at the output of the drift space
becomes
t %(u)er> u » + S g (u)
where D equals the delay in drift space in seconds.
This spectrum is again modified as it passes through the resonator. The
resonator has the effect of a low-pass filter on the phase modulation,
where the filter is the low-pass equivalent of the actual bandpass struc-
ture. It thus consists of a resistor in parallel with a capacitor with values
such that the bandwidth of the low-pass structure is just half that of the
actual bandpass structure. For a Q of 100 at G000 mc, the bandwidth is
about GO mc; the equivalent low-pass bandwidth is 30 mc, from which
the low-pass transmission characteristic is therefore
nco)=— -^, (i6)
I + JUT
with
r = 2Q/coq = 5.3 X 10- 9 second.
The spectrum of the phase modulation at the output of the open loop is
therefore
Y(u)S+(u)er*"> + Y(a*)S g (a>)
«*(») » • 7 ■ ^ &(«) (19)
1620 THE BELL SYSTEM TECHNICAL JOURNAL, NOVEMBER 1961
and under closed loop conditions
S f («) = Y(u)S t (u)<ri»D + F(«)5»(a>) (17)
from which
» (a) - i-KU» &(a) - (18)
Substitution for F(oj) as given in equation (16) permits the excellent
approximation ,
1
Mr + D)
which for the 450A klystron is accurate to within about 0.01 db up to
the top baseband frequency of 10 mc. After both sides of (19) are multi-
plied by ju, the following identifications are made :
jajS+iw) = frequency spectrum of f'(t), the first time derivative of
\f/(t), and
S§(a>) = frequency spectrum of d(t).
Since the instantaneous frequency of a wave is defined as the first time
derivative of the instantaneous phase, the time domain equivalent of
(19) can be written as
d(t)
frequency modulation = $ (t) = . -. . (20)
Thus, within the accuracy of the approximation stated above, the closed-
loop frequency modulation is directly proportional to the phase modula-
tion introduced by the modulating signal. Furthermore, the equivalence
of this result to the quasi-stationary result obtained in Appendix B is
possible when the following identifications are made :
_ 2Q = dip
co r dw
oca
Thus, the above result can also be written in the form
86
'm 0(0 „ dv ,v
& It) = tt — V(t)
dw do) dw da
which is the linearized dynamic equivalent of (10).
TERMINAL TRANSMITTER AND RECEIVER 1021
From the preceding analysis the following conclusions are obtained:
(a) Phase modulation is dynamically converted to frequency modula-
tion by regenerative action in the feedback loop.
(b) Klystron modulation sensitivity, inversely proportional to open
loop delay, is the same as that found in the steady-state analysis.
(c) The resultant frequency deviation is essentially independent of
video signal frequency.
APPENDIX D
Klystron Fluctuation noise
A comparison of the FM noise characteristic of 450A klystrons, Fig.
10(d), with noise characteristics typical of other electron tube devices
leads to the conclusion that the two phenomena are identifiable.
"Flicker" noise, varying approximately as 1//, predominates for fre-
quencies less than 25 kc. "Shot" noise, flat with frequency, predominates
at frequencies greater than 25 kc. Since shot noise is controlling over all
of the message band and a major portion of the television band, it is the
more important source of klystron noise.
Flicker noise is associated with random time variations in group elec-
tron emission from the cathode. The power spectrum for these variations
decreases at about a l//rate. The exact mechanism whereby this random
amplitude modulation of the dc space current is converted to frequency
modulation of the carrier has not been quantitatively identified. It
probably comes from variations in the inter-action grid capacitance or
drift space delay induced by the space charge density modulation.
On the other hand, shot noise has been clearly identified as being due
to the statistical time variations for single electron emissionor ( capture
by intervening grids), which have spectral intensities that are relatively
constant up to and above 6000 me. The electron stream flowing through
the resonator interaction grids [twice, as illustrated in Fig. 11(a)]
varies randomly with time. Those frequency components of the varia-
tions which fall in bands equally displaced on either side of the carrier
add to it, and randomly vary its phase. This phase variation is converted
to random frequency deviation by the action of the regenerative loop as
described in the previous section.
Shot noise per cycle of bandwidth is given by the following equation
for temperature-limited emission :
i;- = 2(3.18 X 10- 19 )/ (21)
where i s is the rms fluctuation current and / the dc beam current. The
1G22 THE BELL SYSTEM TECHNICAL JOURNAL, XOVEMBER 1961
factor 2 is used to allow for the double transit of the electron stream
through the interaction grids.
Also flowing into the interaction grid space is a bunched electron
stream which has an equivalent fundamental carrier current component,
i e , at radian frequency, w . This current produces power, P, in the
resonator load. Therefore,
where
R = resistance of the loaded resonator, and
C = capacitance of the interaction grid space.
This carrier current will be randomly phase modulated by upper and
lower sideband noise currents to give a net open-loop rms phase deviation
in each one cps band of
0„ = % = fa \/ -—j, radians . (23)
Regenerative action described in Appendix C will change this to a fre-
quency deviation which is independent of the noise spectrum frequency.
Neglecting D with respect to t in equation (20) :
rms frequency deviation = — = -r^r-
T SiKJ
(24)
= i„ \/ p" rad/sec in a one-cps band due to noise.
Values typical of the 450A klystron are :
27 = 0.1 amp,
P = 0.2 watt,
C = 0..5 micromicrofarad,
Q = 100, and
o>o = 2tt X 6175 X 10 6 .
When substituted in (24), these give an rms frequency deviation in a
one-cps band due to shot noise of 0.9 cps.
Experimentally, it is found that one 450A klystron generates an rms
TERMINAL TRANSMITTER AND RECEIVER 1623
frequency deviation of about 0.-4 cps in a one-cps band*; for the two
klystrons in the FM modulator the noise is 1.41 times greater.
Apparently space-charge smoothing, which usually reduces rms shot
noise in space-charge-limited devices by a factor of 5, gives a 2/1 im-
provement in this klystron. Alternatively, space-charge smoothing may
be more fully effective, and partition capture may be responsible for the
added noise.
APPENDIX E
Discriminator Network Design
To demonstrate analytically the main features of the discriminator
network design, the magnitudes of the interstage impedances for the
common interstage and for the high- and low-frequency discriminator
networks as shown in Fig. 7 are expanded in power series about the
center frequency of the discriminator. For the common interstage this
gives
| Z e | = | Z c0 | (1 + ad + c-,8- + c ;! 5 3 • • •), (25)
where
8 — lo — wo , and
coo = center frequency of the discriminator.
Similarly, for the high- and low-frequency discriminator networks,
\Zi\ = | Zw | (1 + Zi* + lid 2 + l s 5 3 + • • •) (26)
I Z h | = | Z h0 1 (1 + M + W + W + • • • ) • (27)
The amplitude characteristics for transmission through the two sides of
the discriminator are (hen obtained as the product of | Z ( . | with | Z t |
and | Zn | respectively. Thus,
,1/ = ki\Z e \ \Zi | (28)
A h = /;,, \Z e \\Z h \ (29)
when 1 the factors h t and k h are introduced to include the effect of electron
tube gains in the two sides.
For the purpose of the following discussion the constant, Ci , will be
taken equal to zero. This is done in the actual design by tuning the in-
* To generate this same noise by thermal agitation would require a resistance
in the repeller circuit of 5 megohms; the actual circuit value is 800 ohms.
1624 THE BELL SYSTEM TECHNICAL JOURNAL, NOVEMBER 1961
terstage network to the center frequency of the discriminator. Per-
forming the multiplications indicated in (28) and (29) and discarding
terms higher than <5 3 gives
A i = A,o[l + M + (h + ct)8* + (/a + c 3 + /ic 2 )5 3 ] (30)
A h = A M [l + M + {hi + c,)6 2 + (h 3 + c 3 + /iic 2 )5 3 ] (31)
where
A,o = ki | Z e o | | Zio | (32)
AfcO — &/. | ^r0 | I ^/.0 I • (33)
The input signal has the form,
i(t) = i [l + ,4(0] cos M + *(fl] ( 34 )
where
A(t) = amplitude modulation,
,p(t) = phase modulation, and
(p'{f) = frequency modulation.
As a result of the amplitude shape in the discriminator networks, the
instantaneous amplitudes at the output of the networks (or input to
AM detectors) are approximately* as follows:
Mi(t) = ioA lo [l + A(t))
[1 + Uv' + (h + c 2 V 2 + (/ a + c 3 + hc 2 )<p' s ] (35)
M h (t) = ioA M [l + A(t)]
[1 + h\<p' + (h 2 + c 2 V 2 + (/i 3 + c 3 + /mc 2 )^' 3 ] (36)
The output of the balanced discriminator is given by the difference
between the outputs of the two detectors. If the diode detector efficiencies
for the two sides of the discriminator are D t and D h , the discriminator
output is given as
Discriminator output = D h M h (t) - D t Mi(t). (37)
* These results are based on a quasi-stationary approach and are valid for very
low modulating frequencies. At higher modulating frequencies the relationship
between instantaneous frequency and instantaneous amplitude is more complex.
The additional terms which then arise are due primarily to delay distortion which
is compensated by delay equalization (see Section 7.2).
TERMINAL TRANSMITTER AND RECEIVER 1625
The individual output terms are listed and discussed below. For
simplicity in the expressions which have to be written we let Ei =
i a A iqDi and E h = ioAhoD h .
(i) E h — Ei = dc output.
(ii) (Eh — Ei)A(t) = output due to unsuppressed amplitude modula-
tion. By adjusting the relative gains in two sides of the discrimi-
nator so that the dc output is zero, this term is also eliminated.
(iii) (E/Jii — EiU)<p'(t) = desired signal output. Since hi is positive and
h is negative, the signal components from the two sides actually
add in the output.
(iv) [Eh(hi + c 2 ) — Ei(h + C2)]<p'(tY = unwanted second-order mod-
ulation. This term will be discussed in detail later.
(v) [E h (h 3 + c 3 + tiicj - Ei{h + c 3 + hc 2 )]<p\t) 3 = unwanted third-
order modulation.
(vi) Finally there is a set of terms identical to (iii), (iv), and (v) above
except that each is multiplied by A(t). By keeping A(t) small by
means of limiter action ahead of the discriminator, these terms
are held to acceptable levels.
The effect of the common interstage is demonstrated by examination
of the distortion term listed in (iv) above. In the absence of the common
interstage, this term would be
[E h h 2 - Ejojv'ity-
and, with E h = Ei to minimize distortion term (ii), it is desirable, and
possible, to adjust the discriminator so that h* = I. and the term goes
to zero. For the values of h 2 and l-> of the actual discriminator design,
the requirement on the stability of E ti and Ei to meet the modulation
objective would then be given by
I /<• _ F, I < ' E " '
\ I'jh — tbi\ ^
15
This corresponds to holding the relative gains of the two sides of the
discriminator equal to within about 0.5 db between maintenance inter-
vals.
The use of a common interstage selected so that c 2 = — ht — —h
causes the distortion term to go to zero even though Eh and Ei are not
equal. This is illustrated in Fig. 9(a). In practice, it has been found
1626 THE BELL SYSTEM TECHNICAL JOURNAL, NOVEMBER 1961
possible to adjust c 2 such that c 2 + h* = c 2 + h are about 10 per cent
of h 2 = / 2 , which relaxes the requirement on gain stability to
Eu - -Bl | S i r ,
1.5
which corresponds to a relative gain stability of 4.4 db.
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1 Kinzer, J. P., and Laidig, J. F., this issue, ]). 145!).
2. Barstow, J. M., and Christopher, H. N., Trans. A.I.E.E., 72, Part 1, pp. 735-
741, Jan., 1954.
3. Kelly, H. P., Trai .. ..
4 Roetken. A. A., Smith, K. D., and Friis, R. W., The 1D-2 Microwave Radio
[elly. H. P., Trans. A.I.E.E., 73, Part 1, pp. 505-509, Nov., 1954.
oetken. A. A., Smith, K. D., and Friis, R. W., The TD-2 Microwa
Relay System, B.S.T.J., 30, Part II, pp. 1041-1077, Oct., 1951.
5. Haury, P.T., and Fullerton, W. ()., this issue, p. 1495.
(i Sproiil, P. T., and (iriffiths, H. D., this issue, p. 1521.
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